GB2113025A - Thyristor switching circuits - Google Patents

Abstract

The circuits include an SCR or triac (2) having its gate (G) permanently connected to the ac network, via a suitable impedance (3, 4), which impedance admits a gate current near the minimum necessary for original turn-on (for triacs, at least for the I+ mode) at lowest ambient temperature, which current is sinusoidal, and which current is reduced below the minimum needed for turn-on by a control circuit by direct shunting of the gate to a minus terminal (MT1) by a negative- coefficient semiconductor (v.s. heat, light, etc.), or a switch (not shown). Said semiconductor may be a transistor with favorable Iceo characteristics, e.g., the OC171, or a NTC resistor and said switch is usually a semiconductor switch. Capacitive- gate-feed embodiments, where gate current leads anode voltage by essentially 90 DEG el. allow full-wave almost-zero-voltage switching; placing the feed impedance before the load allows quasi-d.c. switching. By simple Ohm's-Law tapping of a resistor between feed impedance and gate, sensitivity is easily increased. By using a transistor amplifier as buffer, small currents can control the thyristor for variegated applications. The relatively low gate currents used facilitate the above, using the simplest components and circuits, and prevent damage to and difficulties with the thyristor. The applications presented are automatic, proportional, or manual, and include water temp. control, and on/off and dimming lighting control. <IMAGE>

Description

SPECIFICATION Automatic proportional and manual electronic power control and switching via normallyclosed thyristors directly sinusoidally gate-fed from anode or MT2-side of AC line via static impedances admitting near-minimum gate current by variable direct gate shunts with options for quasi-zero-voltage and quasi-d.c.

switching Background of the invention The triac and SCR solid-state thyristors are the basic building blocks of state-of-the-art switching and power control, whether for consumer or industrial use. The two main applications of the above are switching, i.e., the turning on or off of the device from one defined state to another; and control, a special form of switching, where the final state to be arrived at may be varied either automatically or manually, to give various outputs, according to varying conditions external to the circuit.

Most forms of switching and control are normally-open (OFF) circuits, where the thyristor at power-up is non-conducting. Then, the gate receives an a.c. or d.c. pulse, supplied by a capacitor (usually via a disc, e.g., in phase control), or a pulse transformer; a d.c. voltage (as from a separate power supply, optionally via a control buffer such as an Op. amp); or, the sinusoidal line voltage, invariably as in common usage, via a suitable series resistor, acting as a simple on-off switch, where, as said resistor is connected from between load and the thyristor MT2/anode terminal, to the gate, gate drive is applied only until the voltage across the thyristor has dropped to its ON value. After a period in the used range, the thyristor turns on, supplying power to the switched or controlled circuit.

The gate current supplied is usually in large excess of that actually needed for turn-on at prevailing conditions, especially when phase control is used. This is done to prevent di/dt damage (see RCA Application Note AN-4242, and "Proceedings of Powercon." 2nd National Power Conversion Conference. Vol. II. Oct., 1975, p. 207), and lower RFI by reducing turn-on time.

The said current is invariably either in phase with, or lagging behind the line voltage by 0--900 el.

(the latter case being the basis for phase control); there are almost no examples of leading gatecurrent feed in common-place applications of thyristors, and those very few that exist are, in reality, hybrids, either of leading sinusoidal gate feed (which may be more correctly called cosinusoidal feed) with pulse control (see, e.g., 'Thyristor Control", p. 89, 1973 (Mazda, ITT); or, a hybrid of sinusoidal resistive-gate-feed with pulse control (e.g., RCA Transistor, Thyristor, 8 Diode Manual, pp. 642-3, 1969). These hybrids give normally-closed integral-cycle control, it is true, but are complex compared to the proposed method, as will be seen below, as will be other advantages and new operations that the proposed method allows.

To turn off the thyristor, and, thereby, the load current (and voltage), the gate current is interrupted: the trivial case of disconnecting the gate feed by mechanical switching is not pertinent here. In the case of a capacitor-diac combination, a sensor may change the phase difference by shunting the capacitor, thereby lower the duty-cycle, or completely stopping the capacitor pulses to the gate, via the diac; a small voltage change to an Op. amp. feeding the thyristor gate can also accomplish the same.

The shorting of the gate directly to the MTl/cathode terminal was found only in a few cases (see SCR Manual, General Electric Corp., pp. 199, 223, 1973), notwithstanding the danger of negative (outflowing) gate current mentioned in this same source (p. 81) (also see Westinghouse SCR Designers Handbook, p. 6-4, para. 6.1.3).

The possibility of shunting the gate (as opposed to shorting it) by a variable (but nonzero) resistance, or, even more usefully, shunting a tap on the feed impedance, above the gate, has not been found in the Art, although it has many advantages. The absence of such a technique probably stems from the usually-large gate currents used, which, as mentioned, are usually much greater than necessary. The use of leading (capacitive) feed, even when it is taken from before the load -MT2/anode-terminal connection (so that it is "quasi-d.c." in that it remains ON for the entire gate turn-on period, as in d.c. gate firing), allows lowest possible gate currents, which are managable, as will be explained.

Summary of the invention The present invention is characterized as a method for electronic power control and switching to a load via a triac or SCR, where the said thyristors are normally-closed (ON), due to the gate being directly fed with sinusoidal a.c.

from the MT2 or anode side of the line, via static impedances, where the said admitted current is near the minimum needed, at lowest ambient temperature (at least for the l+ mode in triacs, in cases where the current that will flow through the thyristor will be of such a degree as to heat the triac enough to later turn on the III- mode). In most cases, the said feed impedance is capacitive, to give a leading gate current, and "quasi zero-voltage switching", the benefits of which regarding di/dt and RFI reduction or elimination are well known. A favored embodiment of the capacitive-feed method is the "quasi-d.c." option, as mentioned above. Thus, the ideal situation where the gate will be fired close to the zero crossing, and the firing current will be applied long after turn-on is easily accomplished.It might be thought, however, in light of the objections to negative gate current cited above, that, for example, in the 2nd quadrant, the gate current will change polarity, becoming negative, while the load current is at its positive peak value, causing damage to the gate.

However, an analysis of the factors involved will show that, in most cases, there is no such danger: (1) the latching-on of a thyristor means that the flow of MT2- or anode current through the gate junction to MT1 or cathode creates a voltage drop across the diode(s) which comprises said junction, thereby presenting a self-sustaining current source to the gate.Since the anode (load) current is much greater than the gate current, at least in the method proposed here, it will negate the incoming negative current; (2) Only when the load current through the device approaches the minimum holding current is there a possibility of reversal of gate currenthowever, at this point the voltage as well as the current are low, so that there will be no dangerous "hot-spots"; (3) Those very few thyristors (e.g., some of the BTX series) whose minimum holding current is less than the minimum gate current, might preferably not be used for such applications;; (4) With the gate-feed capacitor connected from gate (via a safety resistor, as is always necessary to protect both the gate, and any semiconductor connected to it, from switching currents, etc.) to the conjunction of MT2 and load, so that there is no Quasi-d.c. firing, a capacitor of proper value across the gate and MT1 will hold the gate over and add current during the entire turn-on period. This will also give "quasi-d.c." firing.

The final unique characteristic of the proposed method is the manner in which the device is switched or controlled: by the Ohm's-Law shunting of the gate to MT1/cathode, either the direct shunting of the gate to MT1 (through a resistance of minimum value), or of a tap on the feed impedance above the gate to MT1/cathode.

Usually, the feed capacitor is of such value that the value of the resistor between it and the gate may be changed over a wide range without affecting the gate current by more than a few percent, because of the phase difference between the reactive and resistive current: this is very useful in attaining the optimum voltage for any shunting element used to switch and control the device, as will be seen.

It should be absolutely clear that it does not matter whether the shunting sensor or switch (usually, a semiconductor switch) is directly connected between MT1/cathode and gate; or MT1 and tap above gate; or to a bridge (diode), the a.c. side of which is connected to MT1 and said tap (in the case of a triac, a bridge is necessary for polarized devices, e.g., a transistor used as a sensor; a transistor current amplifier); or to said transistor current amplifier: the same principle of simple Ohm's-Law Shunting applies.

Although the various embodiments will be described in detail later, reference to Fig. 1 will give a general picture of some of the most important embodiments and/or basic building blocks for such, based on the proposed method.

The most powerful building blocks for the method are the combination of Fig. 1 a with Fig. 1 g, the latter being a shunt amplifier; and of Fig. 1 a with Fig. 1 h, the latter being a pulse or voltage amplifier or interface. Other important units are the opto-coupler; the Darlington Amplifier (whether using two or three transistors); the regenerative switch (in any of its various forms, e.g., SCR; UJT; SCS, whether in discrete or integrated form); the Darlington SCR, or DARLSCR, which is unique in itself, and which is a three-transistor unit in which one of two NPN transistors is common to the other NPN of a Darlington pair, and, at the same time common to the PNP of a regenerative switch, to give a device with very high input impedance, suitable for timing in conjunction with an RC circuit, where no charge leakage from the capacitor may be tolerated; and with an SCR-like output.

For those embodiments which depend on sensors, the variables sensed include: heat; cold; light; charge (=time); line voltage; load current; low-level current (touch controls). The scope of the proposed method should not be limited to the above, however, as any variable which can be transduced to resistance, current, or voltage can be sensed by the method in one of its embodiments, and can switch or control electric power.

Besides the common NTC thermistor for heat sensing, the use of the Critical Temperature Resistor is indicated in many applications. Also, it was discovered that the Or171 Ge transistor has favorable properties in the lceo mode, with what seems to be a transition temperature at about 720C; by proper insulation from water being heated in a container, the water's temperature may reach boiling, whereupon the Or 1 71 will shut off the current to the heater, to give a waterboiling control. For cooling control, the common PTC thermistor was used as a shunt. The LDR served as a light detector.

The well-known RC circuit served as a timing element, in conjunction with a regenerative switch; Darlington; or DARLSCR (see above). A resistive-gate-feed circuit (this is the sole embodiment where a capacitive gate feed can not be used) in conjunction with a timing circuit with special voltage source gave a light dimmer/brightener.

Load overcurrent, and over- and undervoltage was controlled. Current was converted to voltage by passage through a low-value resistor, which latter voltage passed through various diodes, or was compared with a standard via an Op. amp.

Voltage was divided and the lower voltage also passed through diodes.

Important switches used were the 555 timer, in various astables, for manual, and automatic regular and proportional controls; and, the 7490, which led to exact power division, which enables a time base, divided into exact fractions, to control the duty cycle of the thyristor. Thus, it is possible without phase control, with a relatively simple circuit, to control lighting or motors. This is especially true for 400 hz power.

Finally, various solid-state relays are shown, where the original circuit is made normally-open, and then this function is negated by simple means, to close the circuit.

Brief description of the drawings Fig. 1 General outline of preferred embodiments.

Fig. 2 Water-boiling control with back-to-back sensors.

Fig. 3 Water-boiling control with resistive gate feed and various sensor connections.

Fig. 4 Heating controls with sensor at low voltage: a) miniature hothouse; (b) opticallyisolated control; c) relay-isolated control.

Fig.5 8 Fig. 6 Heating controls with reset.

Fig. 7 8 Fig. 8 Manual variable-duty-cycle and proportional power controls.

Fig. 9 Exact power division; Fig. 9d Exact lowinertia-load control.

Fig. 10 Lighting controls.

Fig. 11 Cooling controls.

Fig. 12 8 Fig. 13 Timers; Fig 1 3b Automatic light dimmer.

Fig. 14 Touch controls.

Fig. 1 5 Solid-state relays.

Fig. 1 6a-b Overvoltage relays; c-d Missingphase (phase-shift) relay/low-voltage ("brownout") relay.

Fig. 1 7 Solid-state fuses.

Description of the preferred embodiments Referring to Figs. 2; 3; 5 and 6, various nonreset or reset heating controls, in which the sensor is at mains potential (either phase, or, preferably, neutral), are shown. Referring first to Fig. 2a, the preferred capacitive-gate-feed embodiment is shown. Slightly after the a.c.

voltage zero crossing, the leading current through load 1 and feed capacitor 3, essentially at its peak value, passes through protective resistor 4 and, when the sensors are far from the turn-off point desired, passes in sufficient amplitude through the gate of triac 2 to fire latter, whence the forward voltage across same drops, and the mains voltage is across the load, until almost 1 800 el., when either the mains voltage or the load current drops to a value below that necessary to maintain conduction. As known, the gate current in such an arrangement, where the feed impedance is between load and MT2, is cut off after the thyristor has fired, and the forward voltage is established.Very soon after the 1 800 crossing, the gate current, which is now negative, and at its peak will fire the triac in the opposite mode, viz, the III- mode (as opposed to the previous l+ mode). As the heater load heats the medium desired, the sensors in some kind of thermal contact with said medium, will heat up, either almost immediately, or, after a delay, depending on the time constant of the sensor and the insulation existing.After a certain time, depending upon the rate of heat addition (a function of load power, medium volume, and medium heat capacity), the sensor(s) will reach a temperature so that (for a NTC-negative temperature coefficient device) the resistance will have dropped to such a value that the shunting of the gate current, a function of the specific properties of that sensor, will be so great as to take a proportion of the feed current, according to Ohm's Law, so that the gate can not fire at any value of triac blocking voltage. Thus, the heating will be curtailed, until that time when the cooling off again of the sensors to below the abovementioned turn-off point, will allow the device to turn on once more. This process will repeat as long as there remains material to be heated.It should be noted that, in the control of boiling of water, the water itself controls the temperature, as, at STP, the temperature remains at 1000C as long as water remains in the liquid state. The same is true for change of state of any substance.

Thus, it is not necessary to employ a sensor that will turn off the triac at exactly 1000C; all that is needed is to delay the arrival of heat to the said sensor, by proper insulation, until enough heat has been added to the water to boil it. (For more exact applications, the thermal lag must be minimalized.) Figl 2b shows one possible way of adapting a polar sensor (the PNP Ge Or171 transistor) to the a.c. gate current, where the transistors 8 and 8a are back-to-back (anti-parallel), so that, as connected to the triac in Fig. 2a, 1+mode current flows through transistor 8, while Ill--mode current flows through 8a, both via sensorprotector resistor 21. Other possibilities are shown in Fig. la together with Figs. 1e and If; and in Figs. 3c and 3d.In this example, whatever the sensor use may be, it should be especially noted that, as the gate current is never more than about 3.3 ma, and the gate voltage is about 2 volts, and that both the gate voltage and the value of Vc6o decrease with temperature, the sensor self-dissipation will be quite low, preventing both sensor damage, and inaccuracy. This is another advantage of using minimum gate feed. When bidirectional sensors such as thermistors are used, of which the Critical Temperature Resistor (Fig. 2c) is a special example, interfacing is extremely simple.

Sensors NTC sensors which may be used in the circuits related to Figs. 1 a and 1 c or id include thermistors; transistors, especially in the lceo mode, but also in the leCo mode, and especially Ge transistors, because of their greater absolute leakage, and temperature coefficient; and diodes in reverse direction, especially Ge diodes, for the same reasons; this also includes loco, the reverse collector-base current, where the transistor with open emitter is considered a diode; later, it will be shown that such "diodes" have use for temperature compensation.

Thermistors have the advantage of being bidirectional, but have smaller temperature coefficients than either transistors or diodes, except for the Critical Temperature Resistor, which is exceptionally useful for such applications, having a coefficient, at the transition region, of -80%/0C. Such a rapid change is especially beneficial in the case that resistive gate feed is used, for then, the device turns on, after the sensor cools off, at the peak voltage, unless the sensor resistance rises so abruptly that all the gate current is available for turn-on. In any case, it is desirabie to limit any intermediate stage to the minimum, because of "hot-spots" and RFI.

In any case, since only NTC thermistors are available for temperatures of at least 1100 C, they must be used for control at such high temperatures, and in such cases, capacitive gate feed should be used, or the use of an inverted zener series pair is indicated across the gate and MT1/cathode, as will be seen below.

When used as temperature sensors, (Ge) transistors usually are used to give a changing voltage output, which then is amplified to amplitudes capable of controlling the power output (see, for example, Electronics. Engineering Ed., Mar.28, 1958, pp.81-3 and Electronic Engineering, June, 1961, pp. 360-3; however, some circuits are known wherein the lCeo mode is used, although in a different application (see Instrument Construction, Oct. 10,1963, pp. 1- 3).

What is unique about the use of the Or 171 as temperature sensor is that it was found that there is a transition temperature of 73+2 C where the sensor resistance drops about a decade, and that the said resistance values above and below said transition make the Or 171 suitable for controlling heating according to the proposed method, in a precise way. As mentioned, in water boiling controls, there would be a thermal delay to enable the water to begin boiling before the sensor reaches the above temperature. It is believed that this phenomena is "punch-through" (see RCA Transistor, Thyristor, a Diode Manual, 1969, p. 19), although it may be tunneling, or avalanche breakdown.

The following Ge transistors have been found to have useful values of lceo for control purposes (in order of decreasing lceo=l,eakxhFE): h,,): AC1 88, AC153; 2SD30; 2SB 178; 2SB 187; this list is not meant to limit the possible use of other devices for control, but is an example. It is possible, as mentioned, to use a transistor in the leco mode also. Also, a number of sensors may be connected in parallel, series, etc, as needed. This would also make possible obtaining an average value of the monitored temperature; a weighted average may be obtained using different sensor types at different positions in the medium. It is easy to failsafe a control circuit by using two, or more, parallel sensors.A Ge diode such as the 1 N60 has a good temperature coefficient, but must be interfaced with the gate as may be seen by referring to Fig. 3d.

It should be noted that, in many cases, as the sensor is heated, it will at first shut off the IIImode (for a triac), as that mode has higher current demands than the l+ mode; then, the 1+ mode will also shut off, upon further heating (except for the abrupt sensors, e.g., the Critical Temperature Resistor-CTR, etc.); upon cooling, the reference will happen, but, as the triac has also cooled off, the All mode has greater demands, and may not come on at all, until the device shuts off completely once more, thus, the final high temperature will generally be lower than the original high (boiling, for example).

In general, the medium temperature at thyristor turn-off is dependent upon: rate of heating; sensor insulation; sensor position (water at vessel bottom is somewhat cooler than at top); sensor series resistor (e.g., resistors 21 and 27 of Fig. 3b) (no.forthe CTR, etc); base bias (see Fig.

3c), which may entirely override, or partially change, the sensing function (included in this would be the connection of some resistance between base and collector of sensor transistor).

It is obvious that water heating control is only one of the many possible uses of a circuit, other uses being to control space heaters, such as room heaters, floor warmers, hothouses, etc.; electric ovens and stoves; electric irons; protection of heat-sensitive materials or installations, including semiconductor devices; etc.

Referring to Fig. 3, a heating control with resistive gate feed, without reset is shown. Again the sensor(s) is at mains potential, so that sufficient isolation is needed. The operation is similar to the control of the previous Fig. 2, except that the gate will fire later in each mode than for the capacitive-feed embodiment: for example, if the gate needs, at 250C (at power-up) 1 ma (I+), and 2.2 ma (III--), it will fire, very approximately, at 1 80 and 2230 el., respectively (1GT(min) is, as is known, also a function of forward blocking voltages). Since RFI is possible, the filter consisting of 24 and 23 is needed. The sensors are connected as before (Fig. 3b), or through an amplifier/buffer (Figs. 3c, 3d), as mentioned.

The zeners 22, 22a are dimensioned such that, at least the l+ mode wil come on at power-up, at the lowest ambient temperature found. For example, if the zeners are 5.1 v, and resistor 4c is 750 ohms, mode l+ will come on at about 100 v, and, at about 130 v, zener 22 will conduct enough current to lower the gate current below IGT(mjn, for mode l+ (which remains on as the gate has already fired), while zener 22a will keep the gate off when the voltage reaches 220 v, and higher. However, as the triac heats up because of 1+mode current, the gate will eventually fire in the Ill--mode, at 130 v, and below. As the medium, and sensor(s) heat up, the reverse will occur, and Ill-will turn off at 130 v, and later l+ also. When the sensor(s) cools once more, the 1+ mode will turn on at 130 v, but the All mode may not turn on again at all until shut-off, as explained regarding capacitive gate feed.

As can be understood, the above scheme will permit Ill--mode current only if the load current is enough to heat up the triac enough so that the gate will fire before zener 22a conducts enough to shunt off the current, and this will also be a function of the heat-sinking, and ambient temperature.

It is obvious that a SCR could be used in such a circuit, either with a bridge, for full-wave service, or in a half-wave capacity. Only one zener would be needed for the gate threshold control, of course.

Since the RFI is proportional to the power dissipated at turn-on, a reduction of the maximum turn-on voltage from 220 v to 130 v leads to about 2/3 less RFI for the III- mode. It should be mentioned that there is always some RFI even in the capacitive-gate-feed embodiments, for the following reasons: a) since IGT is a function of forward blocking voltage, and since, as the gate current leads said voltage by 900 el., the former will decrease as the latter increases from zero, so that a range of IGT values and corresponding forward voltage exist, where it is possible to fire the device; b) as the device heats up, IGT(m,n} decreases, so that the device can fire at a higher forward voltage; c) during rise time, trt forward voltage falls linearly as forward current rises, so that turn-on is non-ideal (this is not true for quasid.c. firing, mentioned before).

The above-mentioned range of firing values is beneficial, however, for partially-inductive loads, where there is a lag in reaching the value of forward current necessary for latching. In such a case, the gate-current lead of 900 el., for capacitive feed, can deliberately be reduced, e.g., by increasing the value of the protective resistor, to correct for such a lag. As mentioned, said resistance can vary within a wide range without affecting the resulting gate feed currrent importantly.

In parallel with the proposed method for power control and switching, it is also proposed to manufacture the following related integrated circuits, based on thyristors: 1) thyristor containing integral zener(s); 2) thyristor containing integral feed resistor and tap(s)-this would have the added advantage of warming up the thyristor during the standby/OFF periods, which would be useful in colder climates, and also lower the IGT needed on initial turn-on; 3) thyristor containing integral protective shunt, across gate MT 1/cathode, via small resistor. This would necessitate combining techniques for silicon and Ge, or other materials, but should be feasible.

Such a thyristor would switch off upon device overheating.

Referring to Figs. 5 and 6, various schemes for adding a RESET option to heating controls are shown. Fig. 5 shows a reset circuit based on the discovery that, while capacitive gate feed may be used with SCR's also (either full- or half-wave), if the capacitor value is near the minimum needed for initial turn-on, and/or if there is no alternative path for the capacitor to partially discharge, that, once the SCR is turned off, as by gate shunting (e.g. by a sensor), the device will stay off at the next half-cycle, since the capacitor is charged to the point that insufficient gate current will flow.

Only by "resetting" the device, by removing charge from the capacitor and/or adding current to the gate junction, will the device start conducting again (after said shunt is removed also).

Fig. 6 shows reset schemes based on the fact that the heated thyristor has reduced gate current and voltage demands.

Referring again to Fig. 5, the SCR 2 is fed by the bridge 35, for full-wave service. Gate-feed capacitor 3 feeds gate via safety resistance 4, and, optionally, through PTC fail-safe sensor 7a.

Also connected across the gate are an OC171 (or other sensor) for water-boiling control, and arbitrarily, a separate sensor to turn off device when there is no water, initially, or after boiling.

The same sensor may serve both functions, and, optionally, the other NTC may be a fail-safe shutoff device, or may be omitted, as may be the PTC thermistor. At turn-on the gate is fired by the peak current from capacitor 3, and the device remains on until the sensor 8 reaches the temperature where the gate current drops below the minimum needed for turn-on, as above, SCR 2 is a high dv/dt type, and, together with gate capacitor 41, make the device largely immune to turn-on by transients. Even after the sensor cools off, the device will remain OFF because of the charge on capacitor 3, as mentioned. by momentarily depressing N.O. pushbutton switch 39, reset resistor 4a is connected to discharge capacitor 3, and the device switches on, and remains so until a sensor turns it off.The sensor need not necessarily be in the medium sensed: for example, the sensor in an electric kettle or samovar may receive heat from steam generated by boiling water, or may be placed on the underside of the kettle, etc. While there are mechanical devices (bi-metal) which perform the same function, they are subject to malfunction, especially because of hard-water deposits on them. It is not known whether electronic circuits for this function already are available. For large vessels, where the rate of heating may be insufficient to boil the water until the sensor is heated, it may be desirable to add a MANUAL function, whereby the water is first boiled without the control of the sensor; then, the sensor would be switched in, and would keep the water within a certain temperature range.Such a function could simply be obtained by switching in or out the collector lead of transistor 12, for example, the fail-safe functions, if they exist, would always remain connected.

Among devices that may be controlled by such a circuit are immersion heaters, and a bodytemperature medical water heater. In the latter case, as in all cases where accuracy is necessary, and medium temperature permits, the thyristor, on a large heat sink, should be immersed in the medium controlled (although galvanically isolated from it), with minimal thermal delay.

Thus, the almost constant ambient temperature will help define 1GT (It will be necessary to derate the device due to the higher ambient.) Referring again to Fig. 6, MT2, gate, and MT1 are connected to the corresponding points; in the embodiments of Figs. 6bd, respectively, while corresponding points of Fig. 6e go to MT2 and MT1, respectively; and corresponding points of Fig.

6f go to gate and MT1, respectively. In all cases, there is not enough gate current for turn-on because of the various shunts present (for triacs, both modes are sufficiently shunted).

In the embodiments of Figs. 6b and f, the shunts are temporarily removed by depressing the corresponding N.C. switches, 6, allowing device to turn on, and thereby heating the triac enough to sustain conduction when the switch returns to the N.C. position; the l+ mode will come on first, and then the Ill- mode. In the embodiment shown in Figs. 6case, the shunt remains connected, and the device is turned on by momentarily increasing gate current, which heats the device, or, by directly heating the device (Fig. 6e), as with an integral resistor, mentioned above. It is obvious that the thyristor can be a SCR also.

The reason for the use of different types of diodes used to shunt the triac gate in Fig. 6f is because of the different characteristics of each mode. The resistor 44 hastens the turn-on time.

Until the advent of thyristors with integral heat-up resistors, Fig. 6f represents the favored embodiment, as it is faster than the others of Fig.

6, and more trustworthy. The embodiment of Fig.

5, however, turns on immediately. All the embodiments of Fig 6 are applicable only to loads which are large enough to sufficiently heat the thyristors Referring to Fig. 11, various cooling controls are shown. Since cooling is the inverse of heating, the principle is the same as for the heating controls mentioned, except that a PTC thermistor is the sensor. Fig. 11 a shows a full-wave SCR circuit. As mentioned, in non-reset capacitivegate-feed SCR circuits a discharge resistor, 55, and an above-minimum value capacitor 3, is used (the maximum gate current needed for this SCR is 200 ua, while the peak gate current here is 470 ua). Thus, after the sensor has once again warmed up from the medium, the device will automatically come on.Fig. 11 b shows a triac version, which, has the unique mode-equalizing resistor, 44, which preferentially shunts the l+ mode, so that both modes will shut off near each other In case of relay "chatter", the arrangement of Fig. 11 c might be used: 76 is a PTC thermistor with large thermal constant, while the constant of 77 is rather small. At power-up, PTC 7 (Fig. 11 b) is not cold, so its resistance is high. PTC's 76 and 77 are below their switching temperatures, ts, so they have low resistance, and hold the load off until they heat up. As the medium cools, PTC 7 reaches the switching region, and turns the load off for a short period, depending upon the equilibrium with the surroundings.PTC 77 cools off; if, for some reason, the triac switches on again, PTC 77 will receive the current, since its cold resistance is very low. Later, both PTC's 76 and 77 will cool off. When the medium warms up, and PTC 7 will allow the triac to turn on, PTC's 76 and 77 will take the initial current, holding the load off; because of PTC 76's larger time constant, it will hold the load (relay) off until the triac is positively on. For loads with compressors, a reset device which delays the turn-on by a set time may be used. For relays, and other inductive loads, a RC snubber circuit across the thyristor (or the load) should be used.

The above cooling controls may be used to control the turn-on of a cooling fan.

Referring to Fig. 4, various heating controls with the sensor at safe voltages are described. It is isolated from the mains by either a transformer, relay, or opto-coupler.

Fig. 4a shows the circuit for a low-power control used for a miniature hothouse, with resistive feed. As mentioned above, change of the feed impedance will change the shut-off point (when using non-abrupt sensors). Thus, as potentiometer 4a varied from 37k to 47k, the final temperature attained varied between 30.8 and 30.00 C, in the apparatus to be described below. With pot 4a at 37k, the device did not shut off, but the duty cycle, after the hothouse Interior warmed up, cycled between two values; when the setting was 47k, the duty cycle varied between zero and some value lower than the above. Such variations could be observed visually from the LED 26a. Capacitor 28 made conduction possible at lower voltages, by raising the load current above the holding current.Thus, with said capacitor, blinking was noticed before shut-off, while, without it, the device shut off immediately, at the same pot. setting.

Referring now to Fig. 4b, an opto-couplerisolated heating control is shown, which uses "quasi-d.c." capacitive gate feed. The leading current flowing via impedance tap resistor 5 and the gate causes a voltage drop of about 20 v (peak) at the junction of said resistor and resistor 4. The bridge, connected with a.c. input across said tap, will conduct when the bridge output is closed, and shunt the gate, cutting off the thyristor. When the sensor 7/8 is cool, the resistance is high, and the current passing from the smoothed output of the transformer-powered d.c. supply is insufficient to cause the LED input of opto-coupler 12 to emit enough light to allow the output transistor (photoresistor) to shunt the bridge output until the triac is cut off. When the sensor resistance drops, the triac will be cut off, and remain so until sensor cools sufficiently.

The current-limiting resistor 32 may be varied to change the cut-off temperature (for non-abrupt sensors). The load may be a relay, with the proper snubber circuit.

Fig. 4c shows a commercially available waterheater protector (Model NFH-1, Oz-On Ltd.), to which the addition of components 4a, 7/8, 16, 2 and 3 add to the original non-water-fail-safe function, a boiling-control function. Optionally, switch 6 makes possibe choice between NORMAL, CONTROLLED, and OFF operation, while switch 39 gives a reset option. In the operation of the now-available device, the relay 1 a is connected directly across the bridge output and capacitor 36. When electrode 38 is in contact with water in the protected vessel, the water completes the a.c. circuit via the grounded metal vessel case and via the bridge 35 input, thereby allowing the relay to supply current to the heater coils. As soon as the water has boiled out enough to break the contact with said electrode, the relay drops out.In the proposed addition, the current flow in the relay is via the SCR 2 in series with it and the negative terminal. The SCR, in turn, is controlled by the sensor, 7/8, as in many previous examples. The gate feed is about 390 el.

ahead of voltage; the capacitor 1 6 also helps to adjust for the partially inductive load. When switch 6 is in the top position, the gate is not shunted, and operation is normal, the heating being controlled only by the presence or absence of water. In the middle position, the sensor is connected to the gate and cathode, and controls boiling, as long as the electrode is in contact with water in the vessel. In the bottom position, the gate is always shunted by the low-value resistor 6a, and the device remains off. With resistor 4a and capacitor 3 permanently connected in parallel, the device will automatically come on after the sensor 7/8 cools off; using the N.O.

switch 39, the device will remain off after the sensor cools off, because of the charge on the said capacitor, as explained earlier for reset controls. Momentarily depressing the switch 39 will fire the device, which will then stay on. The value of capacitor 3 may have to be varied somewhat to fit the SCR gate demands. An audio signalling device, such as a beeper, may be added across the SCR to signal boiling and device turnoff.

Returning to the miniature hothouse, following is a description of one possible physical embodiment of same.

The outer housing consists of a box of dimensions 31 x 17 x 10 cm3. The heating element is a rectangular arrangement of dimensions 20x7 cm2, symmetrically suspended from the removable box cover, the center of which array is also the cover center. Also suspended from the center of the cover is the sensor, the lower edge of which is 4.5 cm below the cover surface, just touching the earth in the lower box, to be later described. The heating element is a few mm above this depth; however, their depth may be varied so as to protect delicate buds from intense heat, if necessary. This is accomplished by mounting their connecting wires on a movable platform. The sensor may also be raised or lowered, if desired, to control the temperature of either the air, or soil, which may differ.As mentioned, by using more than one sensor, an average, or weighted-average reaction can be obtained. The sensor may be painted black for better heat absorption. The inside of the outer housing was painted white for heat reflection.

There are a number of holes about 3 cm from the bottom, for air exchange. The lower box is made of a foam insulating material ("Kalkar"), with dimensions 24x12x8.5 cm3 (o.d.). The wall thickness is about 1.6 cm. Said box was filled with earth, and some water-retaining material, to a depth of 2.5 cm, moistened, and seeds planted in furrows in the soil. Carnation seeds planted out of season, germinated with the aid of the above in high yield, at an accelerated rate.

Referring to Figs. 7 and 8, various variableduty-cycle manual or proportional controls, based on an essentially fixed time base (555 timer used) are shown. Fig. 7a shows the time base for the manual controls, while Fig. 8d shows the time base for the proportional controls. In the latter, the manually-adjusted potentiometer is replaced by the two (or one) thermistors, which perform the same function. Fig. 7a also shows an abbreviated symbol used to represent the timer base: either the manual, or the proportional (i.e., any manual control will become a proportional control by the above simple replacement.) Referring to Fig. 7b, a half-wave SCR control is shown.As in all the controls in Figs. 7 and 8, except that in Fig. 8a, the timer is internally powered, by use of a diode, resistor, zener, and smoothing capacitor, as seen for components 30, 31,52 and 53. (The use of a circuit similar to that in Fig. 11 c, to be discussed later, allows operation without external power supply, and without need for additional components, e.g. 30, 31, 52, 53 here (see Siemens Application Note, Components Group. Single-phase and three-phase control with Triacs and Diacs, p. 27, for a discrete-component astable multivibrator with almost 0-100% duty cycle possibility).) On power-up, the SCR will fire near zero crossing because of the leading gate current, as discussed above.Diode 9a protects the gate from negative voltage, and the operation is half-wave, as mentioned. Capacitor 31 will also charge up, and power the timer circuit. The timers of Figs. 7a; 8d; and 9e are modified-duty-cycle astables. Ra, the capacitor-charging resistor is the resistance from terminal 4 (+) of the 555 to terminal 7, while Rb, the capacitor-discharging resistor, is that resistance from terminal 7 to terminal 6; in said Fig. 7a, for example, R8 consists of resistor 48 and the upper part of pot 49, while Rb consists of the lower half of same pot. When the capacitor is discharging through Rb, which determines the low-output time, the output can sink current (terminal 3) to GND (pin 1).

Thus, at low output, the positive gate current will be diverted from the gate. If the beginning of the low period catches the gate current before the SCR has had a chance to fire, the SCR will remain off, until the 555 output goes high, and the positive gate current commences, afterwards.

Diode 54 ensures that the gate will be fed only from the mains, and not from the positive timer voltage on high output. Since automatic firing of the SCR is desired, when the output of 555 is high, the gate-feed capacitor was made higher than necessary for original gate firing; a discharge resistor was not needed, therefore.

Referring to Fig. 7c, a bridge enables full-wave operation. The functioning is exactly the same as described for the half-wave version; no gateprotect diode is needed in the feed lead; a lowervalue capacitor, plus a discharge resistor 55 are used. Resistor 53 and the zener also help to discharge capacitor 3. The gate tap resistors of Figs. 7b, c must be of such value that the voltage drop produced does not exceed the maximum allowed 555 input voltage.

Referring to Figs. 8a and 8c, the triac counterparts of the above controls are shown. Fig.

8a shows a control with external timer power supply, which greatly simplifies circuitry, while the control of Fig. 8c contains internal p.s., components 30, 31,52, 53. In Figs. 8a-c, MT2, gate, and MT1 are connected to the corresponding points respectively; the triac may be the TAG 98D, etc. Capacitive gate feed without quasi-d.c. feed is used, but the feed capacitor may also be connected before the load, in Figs. 8a, c. In Fig. 8a, the action of the timer 108 is inverted by the buffer transistor, 10: when the 555 output is low, no current flows through the base of 10, and no shunting of gate current occurs; when said output is high, the transistor fully shunts the gate, via the bridge output, shutting off the triac.It is also possible to connect the timer output directly to the bridge, so that, without any inversion occurring, the gate is shunted when the output is low. In such a case, when the timer supply voltage was 5 vdc, the gate tap resistor 5 was 3.9k ohms.

The control of Fig. 8c is the same as that of Fig.

8a mentioned, except that, due to the internal power supply 30,31, 52, 53, galvanic separation of the timer output and gate were necessary.

Therefore, the opto-coupler 12 was used, as shown. Here too, there is an inversion, and a high timer output leads to a turned-off triac.

Referring to Fig. 8b, a method for the direct gate firing by the timer 46, which, although different from the principle of the invention, was brought here because of the relation to the embodiments of these Figures; it is believed to be novel. When the timer 46 is low, the triac is fired in the If and Ill-rnodes; when it is on, the triac receives no gate current at all. Since the duty cycle of the timer is always more than zero, even at minimum potentiometer setting (when used as an automatic control, Fig. 8d, this will surely be the case), it may be desired to be able to completely shut off the device; however, a hotplate load had a temperature of about 400C at minimum pot. setting, so shut-off may not be necessary.Capacitor 31 should be of sufficient value to give minimum RFI (by supplying enough gate current for all duty cycles). Filter capacitor 24 removes much RFI-voltages.

Referring again to Fig. 8d, the proportional modified-duty-cycle time base is further prescribed. The operation is the same as the manual control of Fig. 7a, except that resistors Ra and/or Rb vary as a function of the measured variable (heat). For example, Rat which controls the on-time of the timer, is comprised of resistors 48 and 49a, the latter of which is a NTC thermistor, while Rb is resistor 49b, which may be a PTC thermistor. As the thermistors heat up with the medium, the NTC resistance drops, while the PTC, which controls the off portion of the cycle, grows in resistance (assuming it is above its switching temperature). The result is that the high time is reduced, while the low time is increased.

Thus, the duty cycle will be drastically reduced, while the total period, T, will not change much, relatively. Of course, with said arrangement, it is necessary to invert the output, so that the low signal will shunt the gate current. The NTC may be a critical temperature resistor, an OC171 transistor, etc. If the resistors Ra and Rb are abrupt at the extremes of the temperature range desired, then, at the lower end the duty cycle will be almost 100%, while, at the higher end, it will be close to zero. In all cases, using both a PTC and a NTC gives more sensitivity, less overshoot, and protection at the higher end of the range.The following Table shows results using the NTC thermistor M812, and the PTC thermistor, P270C11 OC th, S t1,s T, 5 Duty cycle 200 9.5 0.5 10.0 95% 30 8 3 11 73 40 5 10 15 33 where th is the high time, t1 is the low time, and T is the total period.

For a comparison of the proposed methods with state-of-the-art methods for automatic and manual controls using astables, reference should be made to Siemens, ibis., loc. cit., and pp. 28- 9.

Referring to Fig. 9, various schemes for exact power division, by exact division of a time base, are shown. Fig. 9a shows the astable time base used for the embodiments of Figs. 9b, c.

Said timer is high most of the period, going low for a short time; this was done because the 555 uses less current in the high state (output disconnected). These negative-going pulses are fed to the divider circuits of the said Figs. The embodiment of Fig. 9d uses the mains frequency, doubled by a bridge, to obtain the time base. In Fig. 9e, the same operation as in Fig. 7a, or 8d is obtained, but the ratio between switch positions is fixed and exact. It should be obvious that the fraction of power obtained, 1/n, is never really 1/n th of the sign-wave voltsxcurrent, since the thyristor starts conducting some small time after the zero crossing; however, for practical purposes, the division may be considered to give an exact fraction of potential power. As will be seen regarding Fig. 9d, it is even possible to correct for the above, if desired.

Referring again to Fig. 9b, the output from the timer of Fig. 9a is fed to the BD (pin 1) and/or the A (pin 14) inputs: in case it is fed to both, outputs such as 2:8 (high to low), and 1:1 may be obtained, from the D (pin 11), and A (pin 12) outputs, respectively. It should be clear that, although the output from the D terminal is a divide-by-ten function for pulses, it is a divide-byfive function versus time; i.e., 20% of the time the device will be high, and will, accordingly, shunt or not shunt the triac, through the buffer circuitry, according to the inversion(s) used (the output might also directly fire the triac, similarly to the firing of the triac by the 555, as mentioned above).Also, when using an output such as a 2:8 output, the shunting is for integral cycles, when, as will be discussed for Fig. 9d, the divider is synchronized to the mains double frequency. For low-inertia loads, such as lamps, such shunting will cause blinking, and only ratios such as 1 :4, obtained as will be shown, should be used, unless the base frequency is high, such as the 400 hz used in aviation. In general, the fact that the human eye can not discern changes occurring in less than about 50 ms is the criterion for such operation.Following is a Table showing different outputs obtained from the 7490 when the BD input is used, and when the C output (pin 8) is connected to the A input (pin 14) (no other input to pin 14): Pin-2 Pin-3 Output Output connec- connec- at D atA tion to tion to (pin 11) (pin 12) pin 12 pin 8 1:6 (hi/b) 2:5 (hi:lo) 12 9 1:5 2:4 11 9 1:4 2:3 9 8 - 1:2 Other possibilities exist: for example, with the same BD input, but with the D output connected to A input, ratios such as 1:4, 1:5, 1:6, and 1:8, exist. When the input is to pin 14 (A input), and with the A output (pin 12) to BD input (pin 1), ratios such as 1:3, 4:6, 2:8, 2:4, and 1:7 are possible.Obviously, by using various combinations, including the use of dividers as the 7492, 7493, etc., an endless variety of possibilities exists. Also, the input frequency may be doubled by various known means, and then divided by an odd integer, to give further possibilities. Simple switching is all that is needed to give a great variety of power fractions from one circuit. The dpdt switch 57a-b of Fig. 9b allows the output to be taken either versus negative or positive, thereby giving the inverse division with extreme ease; thus, if a certain output v.s.

negative is in the ratio of 1:6 (hi:lo) (14.29%), the said inverse will be 6:1, or 85.71% of the time high, etc. Thus, a motor, for example, could be fed an exact power, which is easily varied, which might be important in precision industrial applications, and which is impossible with phase control. Then too, since capacitive gate feed is used, there would be no RFI, another advantage over phase controls.

Referring to Fig. 9c, the 555 timer is used as a frequency divider (see, e.g., Signetics Application Notes, p. 1 55); when the value of the time constant RaC is slightly more than (n-1) Tun/1.1, where Tin=input signal period, a divide-by-n action will occur; said signal input is to pin 2, the trigger pin. The output is then applied, v.s.

negative or positive, using dpdt switch 57a-b, and appropriate buffer, to the thyristor. Of course, the values of Raw and or the timing capacitor, C, may be varied, by known means, to change value of N. Also, the 555 may directly drive the thyristor, as mentioned above. One advantage of this embodiment is the possibility to use the dual timer 556 to provide the functions shown in Figs.

9a and c.

Referring again to Fig. 9d, the embodiment for exact power division, suitable even for low-inertia loads, will be gone into in detail. Mains-frequency a.c. current flows via resistors 60, on the a.c. side, and 61, on the d.c. side of bridge 59, creating a 1 OO-cps pulsating voltage across resistor 61.

Assuming that the voltage is of sufficient value, a current flows via resistor 62 and (optional) diode 63 through the base of transistor 64 and returns through emitter (and pull-down) resistor 65, the said transistor conducts, and the input of the divider is high. Near the end of the half cycle, the voltage across resistor 61 drops below the value which sustains the necessary base current in the transistor, and conduction ceases, and the 7490 input is pulled down, thereby counting one more pulse. Near and after the beginning of the next half cycle, the transistor conducts again, but, as the output goes positive, no pulse is counted.

Only near the end of said half cycle will another count be made. According to the input and other connections, which may be as described for the embodiment of Fig. 9b, the output will change state after a certain number of such negativegoing pulses. As before, dpdt switch 57a-b inverts the output. The output activates the LED of the opto-coupler 12, the photo-transistor output of which is connected, via points A and B, to the circuit of Fig. 1 2b, which is the basis for many controls, as will be seen. Transistor 64 of Fig. 9d may be a single transistor; a Darlington; part of a Schmitt trigger; part of a regenerative switch; etc., as is needed.The value of resistor 62 may be varied to change the point, before the half-cycle end, where the negative-going pulse occurs: thus, after a certain number of counts, the shunting will be removed (opto-coupler off), and the thyristor will come on slightly before the end of the half-cycle, if desired, and then turn off Since the inverse modes, If and 111+ are involved, since in quadrants II and IV leading current flows with capacitive gate feed, it could well be that the current in the gate circuit will be insufficient for this. If necessary, the opto-coupler 12 may be a Darlington type.

Referring again to Fig. 9e, the diode from pin 7 to pin 6 of the timer allows modified astable operation, so that duty cycles of less than 50% are possible. Thus the high:low ratio will be the ratio of Ra/(Ra+Rb), and the accuracy will depend on the resistor accuracy. Optionally, a second diode may be put in series with Rb, with anode to pin 6, to compensate for the first diode voltage drop. Of course, any number of resistors may be used, and switch 49c will define Ra and Rb.

Referring to Fig.10, various lighting controls are shown; Figs. 1 Oa and b are Dust-Dawn controls, while Fig. 1 Oc is a light flasher, or, a cycling relay.

Fig. 1 Oa shows a dusk-dawn control with quasi-d.c. capacitive gate feed, using a LDR as sensor. Resistor 44 and diode 22b are the mode equalizer. By varying the resistor, the difference in time between the extinguishing or coming on of the Ill- v.s. the l+, or l+ v.s. the Ill-, respectively, can be changed, if desired. The use of a LDR to cut off pulses to a triac gate from a charging capacitor via a diac is known. However, here there is no need for a diac, and, because of the kind of gate feed used, there is no need for an RFI filter.

Fig. 1 Ob is an embodiment where greater sensitivity is obtained by the current-amplifier transistor 10, which latter necessitates use of bridge 9. Across the LDR is a delay capacitor, 11 a; this is used to prevent turn-on by momentary darkness as by an object passing over the sensor. It works as follows: at power-up, said capacitor will be discharged and will charge from the bridge via the transistor base, thereby shunting the gate current, keeping the thyristor off. If the sensor is in light, it will conduct enough current via the base to keep the triac off, even if the capacitor charges up afterwards. If the LDR is dark, the capacitor will charge up, and, once the charge is sufficient to limit the current through it and the base, the gate will not be shunted enough for turn-off.Because of the great sensitivity of this embodiment, the LDR must be shielded from the light of the lamp it controls.

Fig. 1 0c describes a circuit which is applicable both as a light blinker, or a cycling relay; transistors 68-9, together with associated capacitors, resistors, and potentiometers are a discrete-component astable multivibrator in which the on-times of each member are separately adjustable by the potentiometers. It is also possible to have a common potentiometer for both bases, when the changing of the wiper position will change duty cycle.

The output of the astable is taken from the collector of the l.h. transistor via resistor 11, to the base of current amplifier transistor 10, which, through limiting resistor 67 shunts the gate current in the circuit of Fig. 1 2b, mentioned above, via points A and B, when said l.h. transistor is non-conducting. Power for the astable is taken from the bridge of the triac. Capacitor 78 holds the astable over when the bridge voltage goes to zero. It can be connected with positive terminal to cathode of diode coming from A, so that the voltage will not be affected by the periodic turning on and off.

Another lighting control, an automatic light dimmer, Fig. 1 3b, will be described after other circuits based on timers are brought, in Figs. 1 2a, c, and d, and 13a and b.

Fig. 1 2a depicts an interval timer in which the All mode turns off before the l+ mode, with option for almost-simultaneous turn-off by mode equalization, for non-inductive loads, based on a RC timing circuit feeding the base of a Darlington pair. The gate feed is quasi-d.c. capacitive; the gate tap feeds a bridge 9, to the output of which is connected hold-over capacitor 78 and the collector and emitter of the Darlington 13. Timing capacitor C (15) is charged via 16, which may be a fixed resistor, e.g., for timers for stairwell lighting, where the on-time is constant; may be a trimmer, for non-frequent changes; or may be a potentiometer, as shown in the Fig.The purpose of diode 11 5 is to prevent discharge on the half cycle that conduction stops (since the negative side of C is supplied from the negative of the bridge 9, there will be current flow only when MT1 (or gate) is positive). Of course, full-cycle charging may be obtained from the positive side of the bridge; however, half-cycle charging give a timing period more than twice as large as with full-charging, with the same components, which has economic value for long timing periods.

In general, it can be shown that the timing period is given by t=Arc 1 +(1/An-1)) where n=E/V A=R/(R+r) where E is the source voltage (minus any diode drops present); v is the final voltage on capacitor C; R is any resistance in parallel with the capacitor C, either leakage resistance; or a resistance purposely put across the capacitor, as will be explained; or the dynamic resistance of the base emitter diode, a function of forward voltage, if appreciable. For the Darlington of the Fig., of course, this is the series resistance of the two base-emitter junctions. rc is the time constant.

The following may be seen from the above formula: 1) For A close to unity and E much greater than v (n large), the formula simplifies to the approximation t=rc/n 2) The product An must be greater than unity; this simply means that if An is less than unity, the voltage across the voltage divider r-R can never reach v; 3) t may be increased by small A and/or small n.

For the halfwave charging, the period is more than twice the period derived from the above formula because the charging voltage, operating for half the time, is not the peak voltage, but an integrated average of the sine voltage, which was found to be about 74% effective; thus, the period is increased by 2/0.74=2.7 times.

For the circuit of the Fig., v for the turn-off of both the l+ and All modes was about 0.55- 0.56 v (X 2, because of the Darlington connection); the actual time required to reach this voltage was somewhat longer than in the simplified formula, since, as the voltage across the base-emitter rises, the dynamic resistance drops, and the value of the constant A becomes small. Thus, the rate of charging of C will slow up towards the end of the timing period, T. It was found that certain switching transistors, e.g., the BSY79, could not be used in this circuit, since they pass a larger base current for the same baseemitter forward voltage, and the value of A dropped so low that v was never reached.In embodiments where switching is used, as in the regenerative switch of Fig. 1 2c, the value of v is only about 0.42 v, because of the much greater sensitivity, and the dynamic resistance is much greater; therefore, the value of A is changed to a much lesser degree, and the BSY79, for example, was found usable, and the same value of T was obtained. In the Darlington circuit, it suffices to use at least one transistor with higher dynamic resistance: thus, by using one BC237A and one BSY79, the circuit performed, although there was a delay in the Ill- off-time.

Switch 80 resets the timing capacitor C by discharging it through the switch-protect resistor 81. The triac turns on immediately, in both modes. Capacitor 79 prevents blinking near the shut-off of modes l+ and Ill-. An advantage of having different periods for said modes is that, for stairwell lighting, for example, the extinguishing of the All mode first serves as a warning that the lights will soon go out completely; thus, if necessary, the reset could be activated again. As mentioned, mode equalizer potentiometer 44 can change the off-time of the l+ mode.

Referring again to Fig. 1 2c, the regenerative switch composed of PNP transistor 18 and NPN transistor 1 7 is connected via protective resistor 84, to points A and B of the circuit of Fig. 1 2b, which has already been mentioned above. Timer capacitor C, 1 5, is connected across the input of the NPN, and is charged via resistance 16, which may be a potentiometer, as shown, or a trimmer, or set resistor, from the positive voltage developed from the current flow through diodes 88a-c; the diodes are supplied from the zener 86 via resistor 87, said zener being supplied via resistor 85 from the bridge connected to points A and B.The use of the above circuit not only gives a constant-voltage source for the timer, but, in the case where there are abnormal temperature conditions, a compensation for same (this is strictly true only when said conditions do not change over the timing period). Since, as is well known, the forward voltage drop of a diode is a function of temperature, v, the final voltage to be obtained on capacitor C, will change, and, with it, the ratio n=E/v. However, since the diodes 88ac also will change their voltage drops with temperature, the ratio n will remain the same (the base-emitter junction of three transistors identical with the NPN used in the regenerative switch may be used as diodes for exact tracking).Of course, the regenerative switch may be controlled by a timing capacitor charging from any other voltage desired: as in Fig. 1 2a; or full-cycle; or from a tap across a resistor across points A and B; etc. As mentioned, when the voltage across capacitor C is about 0.42 v, immediate switching occurs, the gate current is shunted, and the triac turns off; of course, both l+ and Ill-modes turn off together.

Resistor 84 protects the regenerative switch from the sudden discharge of capacitor 78 in Fig. 1 2b.

Upon depressing N.O. pushbutton switch 80, capacitor C is discharged, and the device is thereby reset, whence the thyristor turns on again. A protective resistor may be connected in series with capacitor C and switch 80. Because of the rapid switching which turns both l+ and Ill- modes on or off simultaneously, the circuit is suitable for inductive loads also; including relays-thus, the simple timer with a relay load can control large amounts of power. Later, a regenerative-switch timing control without thyristor will be shown. It should be mentioned that timing capacitor 1 5 also prevents premature turn-on by transients, by acting as a high-pass filter at the NPN base.

It should be noted that capacitor C charges faster towards the end of the timing period because the PNP transistor 18 is now passing appreciable current v.s. that flowing through resistor 16; in the Darlington circuit, as mentioned, there is a slowing down at the end of this period.

Referring to Fig. 12 d, a latching timer based on the Darlington already described is shown.

Resistor 16, fixed or variable, as desired, feeds current to timing capacitor 1 5, from any of the previous mentioned possible sources; said capacitor is across the input of Darlington 1212a, which latter includes a LED; an i.c.

photodarlington also may be used. The Darlington output is connected to points A and B of Fig. 1 2b, as before, via protective resistor 84. When capacitor C reaches a voltage of about 2x0.55 v, the All mode turns off. As the load has no current through it, MT2 is at mains voltage, and a leading current can flow between MT1 and MT2. From 180 270 el. the current flows from MT1 via capacitor 32c, resistor 32a, diode 30a, and resistor 32b to MT2; from 2700--4500 el. the current flows from MT2 via capacitor 31, at first, and after capacitor 31 is charged up, through resistor 32b, LED of opto-coupler 12, resistor 32a, and via capacitor 32c to MT1. The peak LED current coincides with the peak gate-feed current: near zero-crossing. Thus, the gate is kept shunted in both modes; also, the voltage on capacitor C rises somewhat after turn-off. Dpdt switch 8080a resets the device.

Referring to Fig. 1 3a, a non-thyristor interval/delay relay is demonstrated. Although not based on the original invention principle, it uses two inventions relates to the originals: the use of a high-input-impedance device such as a Darlington or regenerative switch for timing, and the use of the DARLSCR, already mentioned. Power for operation of circuitry and standard relay 1 a having N.O. and N.C. contacts is taken from transformer 34, bridge 35, and capacitor 36. The relay is controlled by the regenerative switch 17-18, which has a Darlington input by transistor 17 a.Timing capacitor C is across the Darlington input, and is charged from resistor 16, which may be variable or fixed, by a fixea voltage developed across zener 90, via diode 11 5. Optional capacitor 117 provides an uninterrupted power supply, in case continuation of timing is desired if mains power ceases; it should be much bigger than capacitor C. The diode 11 5 prevents capacitor 11 7 from discharging upon power failure. Capacitor 79 prevents premature DARLSCR turn-on by transients. Diode 91 is a free-wheeling diode.

Dpdt switch 80-80a resets the device.

Depending upon which output switch 11 6 chooses, the relay may be an interval timer (using relay normally-closed output), or a delay relay (using normally-open output); or, separate outlet sockets may be used. The DARLSCR will conduct when the voltage on C is about 0.84 v. The use of a DARLSCR instead of an ordinary regenerative switch was to reduce the discharging of capacitor 117 even further. In general, some modifications may be made in the Darlington or regenerative switch makeup: a resistor may be placed between emitter of input transistor 1 7a and base of output transistor 1 7 to increase input impedance; however, the timing period will then increase.The NPN collector of the regenerative switch may be connected directly to relay 331 , through a small resistor, while the PNP emitter can be connected to same via a large-value resistor.

It is proposed that the DARLSCR, as well as other timing aids based on the regenerative switch, be manufactured as integrated circuits, especially for use in shunting of thyristor gates.

Referring to Fig. 1 3b, an automatic light dimmer/brightener is revealed. Here, quasi-d.c.

resistive gate feed is used via resistors 4 and 5.

The voltage drop across resistor 5 is rectified by bridge 9 and smoothed by capacitor 78, across which Darlington pair 1 3-14 is connected. Said smoothed voltage also supplies zener 86, via resistor 85, which supplies current via potentiometers and/or trimmers 87 b-c, to diodes 88a--c, which develop a constant voltage to feed timing capacitor C, 1 5, across Darlington input, via variable or fixed resistor 1 6.Variable resistors 87bc are of such values that 87c near minimum rotation (maximum resistance) will cause a voltage drop of somewhat less than minimum necessary to cause All mode to turnoff, while, at maximum rotation, will cause a diode voltage drop somewhat more than maximum necessary to turn off both All and l+ modes. If only two diodes, 88a-b, are used, the said resistances 87b-c will be correspondingly reduced, to give the same range of diode voltage drops. Thus, referred to potentiometer 87c, when the wiper is turned clockwise, the timer will dim the lights, while, when c.c.w., the lights will brighten.Operation is as follows: Pot. 87c is first brought to the position that will give the opposite of the desired action-if dimming is desired, the wiper is turned first c.c.w.; then, pot.16, the speed control, is turned to minimum resistance, thus bringing the voltage on timing capacitor C to either of the extreme values in a very short time.

Then, the potentiometer 1 6 is rotated until the desired speed, and potentiometer 87c is rotated to the desired degree of dimming or brightening.

Obviously, 87c may originally be set for some intermediate brightness to start with, and then the rest of the procedure followed. For ON-OFF option, resistors may be switched in and out between resistor 4 and bridge (the timing and current to zener will continue even with the said resistor connected, but sufficient gate shunting will not occur); or, the gate may be shunted by a small resistor connected from between gate and resistor 5, to MT1, which will also not cut off zener current. As before, capacitor 79 stabilizes the operation. Inductor L 23 and capacitor 24 are for filtering purposes, as RFI is created in this unique phase control. Also, the thyristor should be derated to prevent di/dt damage. A large heat sink should be used, to better define leT.

It should be noted that many functions mentioned for individual timing circuits (temperature compensation; uninterrupted power supply; latching-off; etc.) may be interchanged, and incorporated in circuits not shown to have such options. Moreover, because of the great simplicity of the invention principle, many functions may be combined, since, referring for example to Fig. 1 2b, any action which will shunt the points A and B will turn off the device: thus, a timing control may be connected in parallel to a heat control circuit (sensor(s)), so that either boiling, absence of water, or the passing of a preset time interval will shut off the power. Said circuit may be said to have NOR logic.Also, many circuits which are temporary-off may be made permanent-off (latching) by making one of the control transistors into a regenerative switch, by adding a transistor of opposite polarity.

Referring to Fig. 1 4a, a latching touch-to-off control is described. Regenerative switch 1 7-1 8 is connected to points A and B of Fig. 1 2b, and shunts the gate current of the triac in same when it turns on. The switch is so sensitive, that touching the two terminals shown, or even one of them, is sufficient to initiate turn-on. Such a circuit could be useful as a security device to turn off power machinery, etc., in case of emergency, within 10 ms of momentarily touching said contacts.

Resistors 93a-b remove any danger from contact with said contacts, as only a minute current can flow through the body. Resistor 79a and capacitor 79 stabilize the control from transients. The device is reset by momentarily depressing the N.C. switch 80.

Fig. 1 4a is a temporary-off analog of the above.

The connection and operation is the same as above, except that, after removing contact, the triple Darlington stops conducting immediately, allowing the thyristor in Fig. 1 2b, etc., to conduct at the next zero-crossing, or before, the roles of protective resistors 93a-b, and resistor 79a and capacitor 79 are the same as above. Resistor 79a also corrects for transistor leakage current, which is greatly multiplied by the triple Darlington.

Fig. 1 5 demonstrates various solid-state relays based on the invention principle; Figs. 1 5a-c use the circuit of Fig. 1 2b as the basis. Referring to Fig. 1 5a, transistor 10, the base of which is biased via resistor 11, turns the normally-closed circuit of Fig. 1 2b, to which it is connected at points A and B, into a normally-open circuit, as long as there is no input to points E and F, or no contact between points D and E. When, for example, a reed relay, or even a relatively high resistor, connects points D and E, the base current to transistor 10 in inhibited, as the Darlington pair 95a-b turns on; thus, said double inhibition (inhibition of the gate-current shunting) will let the triac turn on.Thus, the load receives current upon an input to a reed relay, etc., which isolates the input from the desired output.

Referring to Fig. 15b, an a.c. input signal is isolated from bridge 96 which feeds inputs E and F, by a transformer. Said a.c. input signal will perform the same double-inhibit function mentioned above. A capacitor may be placed across the d.c. bridge output, so that the said function will continue some time after removal of the a.c. signal.

Fig. 1 Sc shows a logic-compatible relay. 12 is the LED of a opto-coupler, the other half of which is seen in Fig. 15a as transistor 95a, with or without transistor 95b (in case of a Darlington opto-coupler). Said opto-coupler provides isolation between signal input, at points J and K, and the triac circuitry. Resistor 100 and zener 101 limit the input to the opto-coupler, and allow a wide variety of input voltages. Diodes 99a-b assure that only voltages at logic "1", i.e., above 2 volts, will activate the triac. When these conditions are fulfilled, the said LED turns on, and, again, double inhibition occurs.

Fig. 1 Sd demonstrates an extremely simple step-relay. When switch 6 is closed, the gate current is shunted; when open, the triac can turn on. Said switch may be a maintained-contact type, where one switch depression closes the contact, while the next opens it; or, it may be the kind of crossed wiring used between two switches which control house or hall lighting, where, because of said crossed wiring, each switch can independently perform the opposite of what the other switch did previously.

Fig. 1 6 demonstrates various relays, based on the invention principle, which are controlled by the value of the mains voltage. Referring to Fig.

16a, and the basic circuit 12b, this extremely simple overvoltage relay consists of just one zener dioe, connected at points A and B, to circuit 12b.

Under normal mains conditions, an insufficient current will flow in the zener, said current being a function of the voltage drop across resistor 5 (+1%) of Fig. 1 2b. When the mains voltage rises above a certain point, the gate will be shunted enough to cut off the triac conduction. Of course, this action is temporary, lasting as long as the overvoltage lasts. If desired, a variable resistor may be placed between point A and the zener, to vary the turn-off voltage somewhat.

Referring to Fig. 1 6b, mains overvorlage is sensed by the voltage drop across resistor 102a in series with potentiometer 102b, together in parallel with LED of opto-coupler 12, and capacitor 31. Current flows when MT1 is positive with respect to voltage of the main terminal connected to the load (see Fig. 1 2b), via resistors 102a and 102b, and said LED; via resistor 32; through diode 30 to mains. By adjusting pot.

102b a point will be reached where the voltage causes the opto-coupler to cut off the triac.

Capacitor 31 provides both a turn-off delay, so that momentary overvoltage will not turn off the device, and a turn-on delay, which will keep the device off a while after the overvoltage is gone. If desired, a resistor may be connected between the diode (LED) anode and the remainder of the circuit. The circuit may be made to latch permanently off by adding a PNP transistor, to form a DARLSCR, mentioned above. However, as the emitter of the input transistor, and base of the output transistor of a Darlington opto-coupler are not brought out, it would be necessary to use a normal opto-coupler to which is connected a regenerative switch, to form the DARLSCR. It is proposed that opto-couplers with said lead(s) brought out be manufactured. Or, a photoSCR might be used. The total resistance of 102a and 102b should be no more than about 2.7k ohms.

Referring to Figs. 1 6cd, a missing-phase relay is described. N.C. relay 1 a is the load of the circuit in Fig. 1 2b. When the circuit of Fig. 1 6c is connected via points A and B to that of Fig. 1 2b, opto-coupler 12 when turned on will shunt the gate current of the triac, thereby turning it off.

Since the relay load is a normallyclosed relay, the three phase connections R, S, and Twill be connected to outputs 40a-c, respectively, by the respective relay contactors, since the relay is deenergized. The input of opto-coupler 12 is fed by combining the rectified and smoothed voltages developed by currents, flowing through resistances, from R, S, and T, independently, to neutral (ground). If one of the phases is missing, the voltage developed will not suffice to pass the zener 33 (25 volts), and the opto-coupler will be off, and not shunt the gate current; the triac will turn on, and the outputs will be cut off. Examining the triple bridge arrangement 59a-c, it will be seen that the respective capacitors will charge up (31 a-c) only on that half of the cycle when the lower phase R v.s. S; S v.s. T) is positive to the higher phase: for example, if S is positive v.s.

current will preferentially flow from S through lower-left diode through lower-right diode of bridge 59a to R, then via bridge resistor 61 b (in diagonal). Capacitors 31 a-c will not only hold over the opto-coupler during the other half cycle, but will give a time delay, in case of short phase imbalance, before the opto-coupler turns off, and a longer delay, before it turns on again. If desired, the off function can be made latching, by the means mentioned above. Both on- and off-delays can be lengthened by increasing the value of said capacitors. The N.C. relay may have N.O. contacts so that warning lamps come on when the relay pulls in.

The same principal may be used for a lowvoltage relay, where the power will be cut off to the load when the voltage drops to a predetermined level to protect sensitive inductive loads, except that, for the one-phase system, triplication is not necessary, of course.

Referring to Fig.17, various overcurrent relays, or fuses, are delineated. All are based on the action of the voltage developed by load current flowing through a resistor, from thyristor to a.c.

mains, which voltage causes shunting of the gate current to a degree sufficient for turn-off, when the said load current exceeds the prescribed value, about 7A for the components in the present embodiments. All the controls are permanent off (latching). In all cases, the thyristor is short-circuit protected by a fast-blow line fuse.

Referring to Fig. 1 7a, the load current passes through the resistance combination of resistor 104a in parallel with 104c and pot.104b, and the voltage developed when MT1 is negative v.s. MT2 passes through diode 30 and pot. or trimmer 32a, to capacitor 105 and parallel resistor 106, which act as time-delay. Said current continues through resistance 32b to the opto-coupler input, and then to mains. The output transistor of said optocoupler is connected to a PNP transistor to form a regenerative switch, as explained above.

Components 79-79a prevent false triggering by transients. When the load current reaches the prescribed level, the LED light output is sufficient to activate the regenerative switch, and the device is shut off. Potentiometer 104b controls the trip point; potentiometer 32a controls the offdelay. The gate feed is quasi-d.c. capacitive. Reset is by N.C pushbutton switch 80.

Referring to Fig. 1 7b, the same principle for turn-off as in Fig. 5 is used. Only, instead of the heat sensor turning off the device, the baseemitter junction (transistor 10) is used as a voltage sensor for the voltage developed across resistor 104a and parallel combination pot. 1 04b and resistor 104c. Potentiometer 104b sets the strip point, while potentiometer 106 sets the offdelay. Reset, after turn-off, is by N.O. pushbutton switch 39. Capacitor 107 prevents turn-on by transients.

Referring to Fig. 1 7c, an exact electronic fuse is described, in which the voltage developed by load current flowing through a resistance, as above, is compared with a standard voltage with an operational amplifier, and the output operates an opto-coupler which shunts gate current, as above.

Addition of the PNP transistor gives regenerativeswitch action which latches the device off.

Examining the Fig. in detail, load current flows from MT1 via resistor 104 to mains terminal on positive half-cycles. Said voltage, via resistor 111 b and diode 130, is impressed upon the noninverting input of op. amp. 108. Inverting input of same receives voltage from 2-volt zener 110. Any voltage difference causes current to pass through 5.ilk ohm resistor 112; because of the great gain of the open-loop op. amp., as soon as the voltage from resistor 104 is even slightly more positive than the zener voltage, the op. amp. output will go high, turning on LED of opto-coupler 12. The function after the said LED turn-on is as in previous cases. Resistance 112 may be variable, to set trip sensitivity; resistance 109 may also be variable, to set trip point.Diode 30, resistor 53, zener(s) 52a--b, and capacitor 31 are the power supply for the op. amp. and zener 110. Reset is by N.C. pushbutton switch 80. The maximum reaction time of this circuit, as of circuit in Fig.

1 7a, is 20 ms, while of the circuit in Fig. 1 7b it is 10 ms. It may be desired to connect a resistor between cathode of diode 130 of Fig. 1 7c and (neutral) mains terminal, to better define the said diode voltage drop.

It may be argued regarding the circuit of Fig.

17a, and other similar circuits, that factors such as opto-coupler-LED degradation with time, and/or change in ambient temperature, which will change triac gate requirements, would lead to large errors in determining the exact turn-off point: however, if the necessary opto-coupler current were to be increased even by a factor of two, in order to either turn on the regenerative switch of said Fig., or for example, to shunt the bridge at the thyristor gate, as in, for example, Fig.

16b, the voltage across said opto-coupler input LED would have to increase by only a very small amount. The degradation in opto-coupler LED performance can be corrected over the long-time period by changing the setting of the trip-point controls, as, for example, resistor 104b in Fig. 1 7a.

The above embodiments should not be construed to limit the scope of the invention, which will be defined by the following claims. For example, many combinations of functions showed are possible. Also, resistive gate feed may be used in most cases, and means for correcting for RFI will be needed.

Claims (96)

Claims

1. A switching or control circuit for controlling or switching the flow of electric current from an essentially-sinusoidal alternating-current source via a normally-closed thyristor to an apparatus load comprising: permanent static-impedance gate-feed means admitting gate current not in great excess of minimum necessary for required initial gate firing of said thyristor; permantent gate-protector means, and gatetap means connecting said gate-feed means to said thyristor gate from MT2/anode side; gate shunting means, whereby gate shunting increases sufficiently, either abruptly or steadily, at switching or control point, to cut off said thyristor load current to degree required; and, interface means connecting said gate shunting means to said gate-tap means.

2. A circuit according to Claim 1 wherein said thyristor may be any one of the group comprising: a triac, SCR; gate-amplifying SCR; and, SCR with bridge means for full-wave operation.

3. A circuit according to Claim 1 wherein said static feed means is a proper resistor, which said gate-protector means is identical with said feed means, and which said tap means may be a tap on said feed means.

4. A circuit according to Claim 1 wherein said static feed means is a proper capacitor, and which said gate-protector means is a series resistor of proper value, which said tap means may be a tap on said gate-protector means, where gate current leads mains voltage by essentially 900 el., and where quasi-zero-voltage switching is attained.

5. A circuit according to Claim 2 wherein the series combination of gate-feed, gate protector, and gate-tap means is connected to said thyristor gate from before said controlled or switched load, where said load is on MT2/anode side of thyristor, and where quasi-d.c. gate firing is attained.

6. A circuit according to Claim 5 wherein said static feed means is a proper capacitor, where gate current leads mains voltage by essentially 900 el., and where quasi-d.c. and quasi-zerovoltage gate firing is attained.

7. A static switching circuit according to Claim 4 or 6 wherein said gate shunting means comprises the series combination of a mechanical switch and a proper low-value resistor, and which said combination is connected directly across gate and MT1/cathode.

8. A circuit according to Claim 4 or 6 for the control or switching of an apparatus load with inductive component wherein said gate current lead is reduced from essentially 900 el. by addition of non-capacitive impedance to gate circuit, and whereby correction for inductive-load current lag is attained.

9. A circuit according to Claim 3 wherein avalanche device means are/is connected from a tap means on said feed means to MTl/cathode respectively, of said thyristor, and whereby quasizero-voltage switching is attained.

10. A circuit according to Claim 1 wherein said apparatus load is an electromechanical relay.

11. A circuit according to Claim 1 wherein said thyristor is bidirectional, wherein initial gate current at ambient conditions is sufficient to turn on gate only in mode with lower current requirements, and where load current suffices to heat thyristor sufficiently so that second mode comes on in reasonable time thereafter.

1 2. In a heating control circuit according to Claim 1 wherein said thyristor is bidirectional, the improvement which comprises: shunting gate, when said gate shunting means is near control point, to degree that mode with higher current requirements does not, according to immediate thyristor temperature, turn on, whereby half-wave operation attained, whereby heating rate reduced, and whereby overshoot decreased.

1 3. An automatic reset circuit according to Claim 1 wherein said cutting off of thyristor load current is automatically reversed upon sufficiently lowering said gate shunting, thereby allowing load current to flow, according to immediate thyristor temperature and shunt conductance.

14. A manual reset circuit according to Claim 1 wherein said cutting off of thyristor or load current at switching or control point is rapid, and for bidirectional thyristors, simultaneous in both modes, and which additionally comprises: latching means, whereby arrival of shunting means at said switching point causes permanent gate current removal; and, reset means, whereby, after said gate shunting reduced, said latching is rapidly negated by external means.

1 5. In a control circuit according to Claim 1 wherein said thyristor is bidirectional, where said shunting increases steadily, and where one mode is normally shunted off before the second, the improvement which comprises: the prejudicial permanent shunting of one of the modes by mode shunt, thereby causing premature shut-off of said mode, and thereby equilizing, or enlarging the difference between, shut-off times of both mddes, where said mode shunt comprises a series dioderesistor combination connected from said gatetap to MT1, or in opposite disposition, depending upon mode prejudiciously shunted, and where said mode-shunt resistor may be fixed or variable.

1 6. In a control circuit according to Claim 2 or 10, the improvement which comprises: the addition of off-aid and on-delay means comprising PTC thermistors with low and high thermal constant, respectively, connected across apparatus load, whereby once thyristor is shunted off, said off-aid means prevents spurious load current flow, and whereby once thyristor refired when shunting is reduced sufficiently, load current is delayed until thyristor is positively on.

1 7. A circuit according to Claim 1 wherein bidirectional semi-conductor means is switched or controlled in only one mode, the second mode always conducting, where said bidirectional semiconductor means is one of the group comprising: a triac; and, the combination of a SCR antiparallel with a diode.

1 8. A circuit according to Claim 1 wherein said gate shunting means is one or both of the group comprising: a variable-sensor; and, a switch means.

19. A circuit according to Claim 18 wherein said variable-sensor means is one or more of the group of semiconductors comprising: abrupt-characteristic heat and cold sensors such as certain high-frequency thin-base Ge transistors such as the OC171; V205 avalanche devices; critical temperature resistor; PTC thermistor; steady-characteristic NTC heat sensors such as Ge transistors; thermistors; reversed-mode Ge diodes such as the 1 N60; OCI 71 before or after switching point; Si transistors and reverse-mode Si diodes in combination with suitable amplifying means; LDR light sensor; and, any variable-sensor comprising variabletransducing means, in combination with suitable amplifying means.

20. A circuit according to Claim 18 wherein said variable-sensor means is one or more of the group comprising: time-sensor, comprising capacitor charging from a non-zero-average current source whereby voltage increases across said capacitor with time, where said capacitor connected across emitter and base of suitable transistor; voltage-sensor, comprising voltage-dropping translating resistor feeding suitable diodejunction means, which said junction means may be one of the group comprising: a transistor baseemitter junction; zener diode; i.r. diode member of opto-electric coupler; current-sensor, comprising current-to-voltage translating resistor feeding suitable diodejunction means, which said junction means may be one of the group comprising: a transistor baseemitter junction; zener diode; i.r. diode member of opto-electric coupler; any variable sensor, comprising variable-tovoltage translation means in combination with an operational amplifier voltage-comparator means; and, low-current sensor, comprising current amplifier, and where the low current sensed is from one sources of the group comprising: current flowing upon touching a contact means; leakage current; non-periodic low-level signal.

21. A circuit according to Claim 18 wherein said switch means is one or more of the group of semiconductor circuits comprising: discrete-component timing circuit, such as astable; modified-astable; bistable; monostable; integrated-circuit timing circuit, such as astable; modified-astabie; bistable; monostable; any astable timing circuit containing variablesensor means, whereby proportional control attained; means-frequency-to-pulse converting means; frequency divider based on integrated circuit counting means, such as member of 7490 family; computer output; and, level-discriminating means, such as Schmitt trigger.

22. A circuit according to Claim 1 8 wherein said switch means is one or more of the group of mechanical switches comprising: relay contacts; and, manually-operated switch.

23. A circuit according to Claim 1 8 wherein any number of properly-connected parallel gate shunting means are combined, whereby a NORlogic circuit is formed such that only the absence of total said sufficient gate shunting will allow thyristor to remain on, and where said total shunting may be caused by one of the group comprising: steady-characteristic gate shunting means; abrupt-characteristic gate shunting means; and, mixed steady- and abruptcharacteristics shunting means.

24. A circuit according to Claim 23 wherein a NOR-logic circuit is formed by proper parallel combination means from number of reset means chosen from one of the group comprising: automatic reset means; manual reset means; and, mixed automatic and manual reset means, where said manual reset means may provide only failsafe protection.

25. A circuit according to Claim 1 8 wherein any number of properly-connected series gate shunting means are combined, wherein a NAND-logic circuit is formed such that only the presence of sufficiently-large admittance of each individual gate shunting means will turn the thyristor off, and where said total shunting may be caused by one of the group comprising: steady-characteristic gate shunting means; abrupt-characteristic gate shunting means; and, mixed steady- and abrupt-characteristic gate shunting means.

26. A circuit according to Claim 19 or 21 wherein said variable-sensor means is one of the group of semiconductor sensors sensitive to the same variable comprising: series sensor combination; parallel sensor combination; and, series-parallel sensor combination, where said sensors are distributed in the medium sensed.

27. In the circuit according to Claim 26, the improvement which comprises: the employment of sensors with characteristics such that, and spacial deployment of said sensors in medium so that, a weighted gate shunting is obtained, in accordance with medium priorities and characteristics.

28. A circuit according to Claim 18 wherein said variable-sensor means is a Ge transistor used in the mode chosen from the following group: lCeo; eco; and, both iceo and I,,,, for bidirectional thyristors.

29. A circuit according to Claim 28 wherein said variable-sensor means is any combination of two or more Ge transistors, whereby bidirectional sensing is attained for bidirectional thyristors.

30. A circuit according to Claim 1 or 23 wherein said gate shunting means is/are directly connected across gate and MT1/cathode, optionally via low-value gate-shunt-protector resistor.

31. A circuit according to Claim 1 wherein said interface means is one or more of the group of means comprising: diode, for unidirectional thyristors; transistor current-amplifier means from said gate-tap means to cathode/MTl, where input said current-amplifier means connected to said gate shunting means; diode bridge from said gate-tap means to MT1 of triac, where d.c. bridge output connected to said gate shunting means; the combination comprising diode bridge from said gate-tap means to MT1 of triac together with transistor current-amplifier means connected across said d.c. bridge output and where input of said current-amplifier means connected to said gate shunting means; and, the combination comprising two parallel transistor current-amplifier means of opposite polarity from said gate-tap means to MT1 of triac, where combined input of said currentamplifier means connected to said gate shunting means.

32. A circuit according to Claim 1 wherein said gate shunting means and said interface means in part comprise a common semiconductor unit, which may be one of the group comprising: Darlington; regenerative switch; DarlingtonSCR; and, opto-electric coupler, and where input said semiconductor unit is part of gate shunting means, and output said semiconductor unit part of interface means.

33. A circuit according to Claim 14 where latching means are provided from an interface or gate shunting means, comprising in part a transistor current amplifier, by addition of opposite-polarity transistor to form regenerative switch, where said transistor current amplifier is chosen from the group comprising: single transistor; output-member of Darlington; and, transistor-member of opto-electric coupler, where said. reset means are provided by external means interrupting said regenerative switch conduction, and where a low-value series safety resistor protects said regenerative switch from momentary current peaks.

34. A separate switching or control unit containing the said circuit of Claim 1 with means connecting to a.c. means; means connecting to load, which said connecting means may be one of the group comprising: socket; plug; and, interlock; and where said gate shunting means may be at a distance from said unit, connected by proper insulated connecting means.

35. A separate switching or control unit according to Claim 10 with means connected to a.c. mains; means connecting relay contacts to main load, which said connecting means may be one of the group comprising: socket, plug; and, interlock, and where gate shunting means may be at a distance from said unit, connected by proper insulated connecting means.

36. A solenoid-valve switching or control circuit according to Claim 1 7 wherein said bidirectional semiconductor means originally conducting in both modes, whereby solenoid originally receives full d.c. power, whereby said gate shunting means is variable-sensor chosen from group comprising: time-sensor; and position-sensing transducer, and whereby power reduced to half-wave d.c. after predetermined variable-change.

37. A manual reset circuit according to Claim 14 wherein the thyristor is a SCR chosen from the group comprising: full-wave, where said latching means comprise use of capacitive gate-feed means having admittance near minimum for initial gate firing, where, once SCR cut off by any gate shunting means, reset not automatic, where said reset means is a reset impedance which momentarily shorts and partially discharges said capacitor, allowing sufficient gate current flow for gate firing, and where protection against spurious dv/dt firing provided by employing suitable SCR, gate-cathode of which is shunted by suitable capacitor.

38. An automatic reset circuit according to Claim 13 wherein the thyristor is a SCR, and where the gate feed is capacitive, and where automatic reset means allowing for sufficient gate current flow for gate firing upon said sufficient lowering of gate shunting is chosen from the group comprising: employing capacitance with admittance in excess of minimum needed for initial turn-on; and, permanent discharge resistor means connected from anode side to cathode.

39. A manual reset circuit according to Claim 32 wherein said common semiconductor unit is a Darlington, input base-emitter junction of which receives increasing voltage from remainder of gate shunting means such that said thyristor load current cut off to degree required, which for bidirectional thyristor said cut-off may be in sequence: Ill- mode, l+ mode, and where external reset means reduce said base-emitter voltage rapidly.

40. A manual reset circuit according to Claim 33 or 39 wherein said reset means momentarily shorts the controlling base-emitter junction, and is one of the group comprising: mechanical switch means in series with low-value safety resistor; low-Vcea transistor receiving sufficient base current to sufficiently lower voltage across said junction, whereby gate shunting reduced; cooled PTC thermistor; and, heated NTC sensor, where spurious dv/dt latching and erratic behavior prevented by proper capacitor shunting proper base-emitter junction, and where hold-over capacitor and hold-over and safety resistor ensure continued latching.

41. A manual reset circuit according to Claim 40 which additionally comprises: touch-reset means, whereby momentarily touching contact means causes leakage current flow via additional current amplifier means, and whereby said amplified leakage current flows via base of said low-Vcea transistor, whereby circuit reset.

42. A manual reset circuit according to Claim 1 wherein gate-feed admittance is of such value that with reset shunting impedance means shunting gate, thyristor turns on only when thyristor temperature significantly above ambient, where said thyristor is fired by temperature raising means, which said temperature raising means is one or more of the group comprising: direct heating of thyristor with heating element; admitting additional current to gate by parallel additional gate-feed means; reducing said reset shunting means; and where said reset shunting means is chosen from the group comprising: diode shunt comprising diode connected across gate-cathode of SCR via suitable low value series impedance, in direction of gate current flow, which said diode having value VT lower than VT of non-heated SCR gate; diode shunt comprising antiparallel diode pair connected across gate-MT1 of triac via suitable low-value series impedance, which diodes each have suitable values of VT v.s.

VT,i+ and VT 111~ of non-heated triac gate; resistor; and, reactance.

43. In a heating control according to Claim 18, the improvement which consists of increasing accuracy and precision of control by means which are chosen from one or more of the group comprising: minimizing thermal lag between sensor and sensed medium; using sensor with small thermal time constant; mixing controlled medium; and, lowering heating rate.

44. In a control circuit according to Claim 18, wherein said variable-sensor means has a steady characteristic, the improvement in defining 1GT' whereby control accuracy and precision are increased, as chosen from one or more of the group comprising: employing proper-characteristic ambienttemperature-compensation sensor means connected across gate-MT1/cathode; employing proper temperature-compensation sensor-means across collector-base of main heatsensor transistor; influencing sensor temperature rise by ambient by proper sensor. positioning; influencing sensor temperature rise by ambient influence on heat conduction means from heat source to said sensor; employing proper-size thyristor heat sink; and, placing thyristor in temperature-controlled medium, which may be identical with controlled medium where said medium temperature permits.

45. In a switching or control circuit according to Claim 19, 20, or 21, the alteration in said switching or control point, switching or control range, and calibration by one or more of the means of the group comprising: changing said variable-sensor means; delaying said variable-sensor arrival to sensed value by one or more of the means comprising: change in said physical-variable-sensor position; thermally insulating said heat or cold sensor; and, electrically delaying voltage increase to said diode-junction means, whereby a time delay is realized; application of override signal to said variablesensor base; and, application of override signal to transistor base of said interface means.

46. In a control circuit according to Claim 1 8, wherein said variable-sensor means has a steady characteristic, the alteration in switching or control point, switching or control range, and calibration by one or more of the means of the group comprising: change in said gate-tap means value; change in base bias to said variable-sensor means; change in base bias to said interface means; change in said variable-sensor means total impedance, by one or more of the means comprising: addition of parallel impedance; addition of series impedance; addition of proper series diode; and addition of proper parallel diode; change in said gate-feed means.

47. A manual reset heating control circuit according to Claim 33 wherein said transistor current amplifier comprises one transistor, where NTC heat sensor means chosen from group comprising: reverse-mode Si diode; mode mode transistor; eco mode transistor; abrupt thermistor; and, steady-characteristic thermistor; where said heat-sensor means connected from positive voltage source to input base of said regenerative switch via current divider means, which said current divider means comprises parallel resistor pair such that heat-sensor means leakage current reduced by flowing via that resistor of said pair connected across baseemitter, to proper level, where, for non-abrupt heat-sensor means, switching action is abrupt, and for said non-abrupt sensor means, switching point may be controlled by varying value of one of said resistor pair.

48. A semi-manual reset heating control circuit according to Claim 37 or 47 wherein a PTC thermistor negates said latching means at its switching temperature, said PTC switching temperature being considerably lower than usual control temperature, whereby circuit reset operation is manual unless medium temperature falls drastically, when circuit resets automatically.

49. An automatic reset heating control circuit according to Claim 37 or 47 wherein a PTC thermistor negates said latching means at its switching temperature, said PTC switching temperature being somewhat lower than cut-off temperature, whereby switching action is abrupt in every case, and thereby circuit resets automatically and medium temperature varies over small range.

50. The use or manufacture of any integrated circuit according to Claim 1, 9, 19, 28, 42 or 47 containing one or more of the group comprising: unidirectional thyristor sensor means; bidirectional thyristor sensor means; thyristor; gate-feed resistor and gate-tap means on said resistor; diode bridge interface means; avalanche device means connected internally to tap on said gate-feed resistor; thyristor overheat protection sensor, internally connected; regenerative switch; Darlington; DarlingtonSCR; touch-reset current-amplifier means; current-divider means; and, internal connection means.

51. A control for bringing a medium to temperature of change in physical aggregation state according to Claim 1 9 wherein said variable-sensor has control point before desired aggregation-change temperature, wherein said variable-sensor suitably thermally insulated from heat or cold source, where medium volume, specific heat, ambient temperature, thermal insulation, heat or cold source power, and said sensor thermal insulation allow portion of medium to change state, where thyristor cut-off in reasonable time after said partial change of state, and where said change of state of medium is one of the group comprising: boiling of water, freezing of water; boiling of a liquid; freezing of a liquid; and, melting of a solid.

52. A temperature control circuit according to Claim 1 8 wherein thermal contact between said variable-sensor and heat or cold source is established by one or more of the group comprising: placing said sensor directly in medium, with proper galvanic isolation, where necessary; placing said sensor in proper thermalinsulating means; placing said sensor on outside of mediumcontaining vessel, with proper attachment means; placing said sensor in proximity with heat or cold source; connecting said sensor by thermal-conductingmaterial means to heat or cold source, where said thermal-conducting material is in controlled medium; placing said sensor so that heated vapor at desired temperature will reach it; placing said sensor on outside of mediumcontaining vessel, with proper thermal-insulating means; conduction, convection, and radiation via air; and, radiation via vacuum.

53. A temperature control circuit according to Claim 1 8 wherein the apparatus load controlled is one of the group comprising: beverage-preparation device chosen from the group comprising: immersion heater; kettle; percolator; and samover; cleaning device chosen from the group comprising: washing machine; and, dishwasher; body-temperature medical water heater; space-heating device, chosen from the group comprising: hot-house; room heater; and, floor warmer; high-temperature device, chosen from the group comprising: electric iron; electric oven; electric hot-plate; gas oven with electric valve means; and, oil burner with electric valve means; refrigerating device; and, any installation sensitive to excess heat, where said temperature control circuit provides protective fail-safe means.

54. A body-temperature medical water heater according to Claim 53 wherein suitable thyristor provided with suitable heat-sink means is immersed via galvanically insulating material in water to be temperature controlled, where main steady-characteristic heat sensor in combination with adjustable calibration impedance means defines desired water temperature, where mixing means assures even water temperature and accurate temperature sensing, and were fail-safe abrupt heat sensor assures heater turn-off at some higher temperature not dangerous to patient.

55. A miniature hothouse according to Claim 53 wherein alternating current source is taken from secondary of step-down transformer means, where thyristor is SCR, where gate-feed means is series combination of potentiometer temperature control and fixed resistor of suitable values, where gate shunting means is Ge transistor directly interfaced with gate-cathode, where heating element load and said sensor may be vertically variable in position, independently, which said load and sensor are suspended from cover of outer housing means, which said outer housing means contains inner container of insulating character, in which said inner container is earth.

56. In a water heating control circuit according to Claim 53, the improvement which comprises: formation of a NOR-logic circuit by addition of nowater fail-safe protection means, where said protection means cuts off thyristor in reasonable time after said thyristor turned on without minimum safe liquid volume in apparatus vessel without damage to heating element means, where said thyristor remains off for minimum time necesary to prevent damage upon subsequent thyristor firing, and where said fail-safe means is chosen from one or more of the group comprising:: using main heat sensor as fail-safe sensor, where provision made for special thermal contact means of said sensor with heat source; using special fail-safe NTC sensor means in proper parallel connection with main heat sensor; using fail-safe PTC sensor to cut off thyristor gate current at low-voltage point on said gatefeed means; use of bimetal switch means to cut off thyristor gate current; use of bimetal switch to cut off load current; and, use of i.r. transmitter means coupled with i.r.

receiver means, whereby absence of sufficient liquid will activate said i.e. receiver, and whereby thyristor gate cuttent will be cut off.

57. In a high-temperature-device control circuit according to Claim 53, the improvement which comprises: formation of a NOR-logic circuit by addition of missing-heat-receiver fail-safe protection means, where absence of device to be heated on hotplate or oven causes heating surface temperature rise, which said temperature rise is transmitted to suitable heat sensor means adjacent to said heating surface, whereby thyristor cut off for necessary minimum period.

58. A NOR-logic home-heating control system according to Claim 53 which comprises: electrically-controlled fuel-burning furnace; weighted main heat sensing means distributed in home with proper connecting means, wherein weighting is according to distance from heatentry port, room-heating priorities, and positionheating priorities; auxiliary outdoor heat sensing means, properly parallel-connected to main sensing means, whereby outdoor temperature will moderate control in proper direction; and, furnace-overheat fail-safe protection means, properly connected to main and auxiliary sensing means.

59. A water-heater protector unit wherein load-supplying normally-open relay is controlled by presence or absence of water in heated vessel, the improvement to which, according to Claim to, comprising: formation of NOR-logic circuit wherein either absence of water or arrival of water at predetermined temperature will cause relay to drop out, and where operation may be chosen from one or more of the group comprising manual reset; automatic reset; continuous; and, off.

60. A water heating control for large-volume beverage-preparation device according to Claim 13 wherein heat sensor control point and heating rate cause premature thyristor cut-off, the improvement to which comprises: addition of manual switch means disconnecting heat sensor, whereby continuous heating possible until desired temperature reached, when said heat sensor is switched in by said manual switch means, whereby thyristor will be cut off until sufficient cooling of water, and whereby control will then cycle on and off according to heat exchange with ambient, water temperature varying over narrow range, according to sensor selected.

61. A Strength Selector Percolator according to Claim 46 wherein water in an electric coffee pot container means is heated by electric heating element means to predetermined temperature, selected by variable selector means, whereby coffee in coffee basket means is brewed to desired strength which said coffee strength may range from mild to extra-strong, where heating rate then reduced to maintain temperature around said predetermined level, where continuous full operation of heating element is possible, and where no-water fail-safe protection means provided.

62. A Strength Selector Percolator according to Claim 61 wherein said thyristor is a triac, where gate-feed means quasi-d.c. capacitive, where variable-selector means chosen from one of the group comprising: suitable single Ge transistor heat sensor connected so that triac l+ mode shunted by said transistor lceo mode, All mode by leco mode, where interface means comprise: antiparallel diodes each connected via suitable series shunt resistor to gate-tap means, where said diodes and said shunt resistors raise switching temperature of said sensor, and varying of said shunt resistors allow calibration of said switching temperature, where said gate-tap means is variable, and chosen from the group comprising: fixed resistors; and, potentiometer, and where said switching temperature falls with higher value of said gatetap resistance; and, suitable single reverse-mode Ge diode connected from collector to base of suitable transistor current-amplifier means, where interface means is diode bridge across variable gate-tap means, which said gate-tap means is chosen from the group comprising: fixed resistors; and, potentiometer, where switching temperature falls with higher value of said gate-tap resistance, and where calibration or switching temperature rises with changing trimmer resistance means across base-emitter of said current-amplifier means; wherein said fail-safe means is identical with said heat sensor means, where said sensor means isolated galvanically and thermally by thin isolating means in contact with both underside of coffee pot via heat conduction means comprising thick copper strip, as well as with bottom of enclosure of said heating element means, which said heating means enclosure bottom protrudes into said underside of said coffee pot, where thickness and nature of said isolating means may vary at each said contact point; and where visual and/or audio indicator means suitably connected across thyristor indicate changes in said thyristor modes' conduction.

63. A Strength Selector Percolator according to Claim 1 8 wherein water in an electric coffee pot container means is heated by electric heating element means to temperature predetermined by variable selector means, where reset is automatic, whereby coffee in coffee basket means is brewed to desired strength which said coffee strength may range from mild to extra-strong, where continuous full operation of heating element is possible, where no-water fail-safe protection means provided, where said variable selector means comprises any number of abrupt NTC heat sensor means with ascending transition temperatures, which said sensors directly interfaced with gate via low-value resistor by selector switch means, said combination controlling at least one mode said thyristor, while other mode may be cut off prematurely to lessen overshoot by early-cut-off sensor means, where one or more said heat sensors have fail-safe function also, and where visual and/or audio indicator means indicates operation status.

64. In a heaticg control circuit according to Claim 12, the further improvement which comprises addition of status-indicator means whereby voltage across load, and each thyristor mode, are indicated, whereby progress of heating process ascertained, and where said indicator means may be chosen from one or more of the group comprising: neon-lamp means across load and across thyristor, whereby functioning of thyristor in mode not het out off indicated by half-lit neon lamps; LED means across load for indicating mode first cut off, and across thyristor for other mode, where each LED may be colored differently; LED means connected by diode bridge means across load, and suitable antiparallel LED means whereby individual modes represented, across thyristor, and where each LED may be of different color; any combination of neon lamps and LED's; and, audio-indicator means suitably connected so that audio signal will occur only when both modes cut off.

65. In a switching or control circuit according to Claim 1 , the improvement wherein the input to said gate shunting means is at low potential v.s.

earth, where any combination may be made of the group comprising: step-down power transformer; mechanical relay; opto-electric coupler; and, low-current coupling transformer.

66. An electronic automatic-reset waterheating control according to Claim 13 wherein water heated to desired temperature and held within essentially narrow range thereafter.

67. An electronic automatic-reset water boiling control according to Claim 66 wherein an electric water heater will turn off a reasonable time after water started boiling, remain off until water temperature falls to some temperature lower than boiling point, turn on again automatically, and turn off again after water temperature reaches after-boiling maximum, said maximum which may be at, or below, boiling.

68. A variable-duty-cycle control according to Claim 21 wherein said switch means is an astable-type multivibrator time-base means wherein variable-duty-cycle output of said timebase means causes periodic shunting of said thyristor gate, said shunting occurring after time base switches, where polarity against which output taken and interface means' nature determine which time-base output will cause shunting, and wherein load current varies accordingly.

69. A manual fixed-period control for highinertia loads according to Claim 68 wherein said duty cycle is varied by one of the means comprising: changing ratio of charging-resistor means to discharging-resistor means, where sum of said resistors is constant, by switching means; changing potentiometer setting, where said setting defines charging and discharging resistors; and, selecting exact ratio of charging to discharging resistor by selector means comprising series combination of exact resistances with switching means, whereby exact power division is attained; and, wherein said load is chosen from group comprising: heating load; light, wherein lightblinker apparatus attained; mechanical relay, wherein cycling-relay apparatus attained; and, valve.

70. An automatic proportional control according to Claim 68 wherein at least one of the said timing resistors is an active-component variable-sensor, whereby duty cycle modified by variable sensed.

71. An automatic proportional temperature control according to Claim 70 the improvements to which comprise: choosing said charging resistor with proper temperature sensitivity and said discharging resistor with opposite sensitivity; and, choosing said timing resistors with abrupt characteristics; whereby total timing period is almost constant; whereby greater temperature sensitivity is attained; and, whereby safety factor introduced at temperature extreme.

72. The exact division of electric power according to Claim 21 wherein an accurate fixedperiod time base, such as full-wave mains frequency, is converted to pulses by pulseconverter means, and said double-mainsfrequency pulses fed to frequency divider, such as of the 7490 family, where varying input to said divider and varying output from said timer by proper switching means gives exact fractions of input frequency, said exact fraction shunting gate of thyristor via said interface means, where said output is essentially in-phase with mains frequency, where said mains frequency is high enough, or said frequency fraction large enough so that low-inertia loads may be controlled by said circuit, where said thyristor output is an exact fraction of maximum, and where said low-inertia loads may be one of the group comprising: lighting; stage lighting, motor control; mechanical control.

73. An interval/delay timing circuit according to claim 21 wherein said switch means is any monostable multivibrator where thyristor shunting occurs with changing time-base output according to time-base output chosen and interface means.

74. A self-powered discrete-component interval timer according to Claim 20 where gatefeed is quasi-d.c., where gate is shunted by highinput-impedance current-transformer means, input of said current-transformer means being connected to timing capacitor, leakage of said timing capacitor having sufficiently-small value, said timing capacitor being fed by current source means whereby increasing d.c. voltage created across it with increasing time passage according to total charge accumulated across said capacitor, where said voltage causes increasing current flow across said emitter-base junction, whereby thyristor gate shunted accordingly until shut off, where total said capacitor leakage and said emitter-base current flow may never exceed current supplied by said current source means until thyristor satisfactorily shut off, where reducing said timer-capacitor voltage by reset means below minimum necessary for desired cutoff resets circuit, and where timing period increased by one or more of the group of means comprising: increasing capacitor value; reducing capacitor charging current; and, increasing leakage across capacitor.

75. A mode-delay interval timer according to Claim 74 where gate-feed is capacitive, where said current-transformer means a Darlington transistor, where said gate shunting increases non-abruptly with time, where for triac thyristor, All mode cut off before l+ mode, where mode shunt means can be employed to preferentially either mode for triac thyristor, where sufficient thyristor heat-sinking is used to define 1GT' and where instant reset is afforded by shunting timing capacitor and capacitor across output baseemitter simultaneously by suitable means.

76. A hailway-lighting timer circuit according to Claim 75, where said thyristor is a triac, wherein after III- mode cut off, lighting falls to half-power, providing a warning of impending darkness, whereafter l+ mode also turns off, and were reset is immediate.

77. An all-load interval timer according to Claim 74 wherein said current transformer means based on regenerative switch, where thyristor cut off is simultaneous for both modes for a bidirectional thyristor, and where reset immediate.

78. An automatic gradual light-dimmer phase control according to Claim 74 wherein gate feed is resistive, where said current-transformer means a Darlington transistor, where said gate shunting increases non-abruptly with time, where zener regulated voltage, variable resistance means, and series diode means provide in combination boundary-voltage means, said boundary-voltage means in combination with source resistor means providing current source means, said current source means feeding timing capacitor across Darlington input, where said boundary-voltage means may be varied from voltage value sufficient to completely cut off said thyristor, in both modes for triac, to voltage value insufficient to cause significant change from maximum output, where changing said voltage values changes light output and direction of change, where changing said source resistor changes rate of light output change, where means for selecting full-on and full-off provided, where proper RFI filter means provided, where proper heat-sink provided to better define IGTT and where series diode means provided some correction for temperature changes.

79. In an interval timer as in Claim 74, the improvement to which may be one or more of the group comprising: addition of series diode in said current source means to limit timing capacitor charge loss during portion of a.c. cycle; addition of UPS option by including fail-safe capacitor as voltage source in case of power failure, where said fail-safe capacitor has value much greater than said timing capacitor, and properly-low leakage value; addition of temperature-compensation means; using a mechanical relay as load, whereby power-handling capability increased; and, using mechanical relay with N.O. and N.C.

contacts, whereby option for interval/delay operation attained.

80. A dusk-dawn lighting control according to Claim 1 9 wherein said variable-sensor means is a LDR, where said gate feed is quasi-d.c. capacitive, where said sensor directly interfaced to gate-tap means, where said gate shunting increases nonabruptly as sun rises, and decreases non-abuptly as sun sets, whereby Ill-mode cuts off before 1+ mode, and l+ mode comes on before III- mode, respectively, and where mode shunt can vary time interval between mode changes.

81. In a dusk-dawn lighting control according to Claim 80, the improvement which comprises: eliminating blinking and false turn-on and turn-off caused by short-term light-level changes by interfacing said sensor via transistor-amplifier collector-base to gate-tap means, using bridge in case of triac, and connecting proper capacitor across said sensor, whereby the delay action prevents spurious response.

82. A touch-to-off switching circuit according to Claim 20 wherein said current amplifier lowcurrent sensor is one of the group comprising: regenerative switch, with suitable contact means connected to same via two or more highvalue safety resistors, with transient-protection means across said emitter-base, where thyristor cut-off immediate and simultaneous in both modes for triac, and where reset immediate by reset switch means;; Darlington transistor of suitable amplification, where gate-feed capacitive, with suitable contact means connected to same via two or more highvalue safety resistors, with transient- and leakage-protection means across said emitter base, where thyristor cut-off may be function of touch-contact pressure, whereby III- mode may cut off before l+ mode, and where reset immediate upon removing finger from contact means; and, Darlington transistor of suitable amplification, where gate-feed resistive, with suitable contact means connected to same via two or more high value safety resistors, with transient- and leakage-protection means across said emitter base, where touch-contact pressure can cause temporary duty-cycle lowering, light dimming, and where reset immediate upon removing finger from contact means.

83. In a touch-to-off switching circuit according to Claim 82, the addition which comprises: delaying reset by reset-delay capacitor means wherein capacitor of sufficient value across interface means will hold thyristor off until it recharges after reset means enactivated.

84. An isolated-input normally-open solidstate relay according to Claim 31 comprising: current-amplifier means properly connected across gate-tap; permanently-connected current source to said current-amplifier means, whereby said normallyclosed thyristor circuit converted to normallyopen circuit; current-amplifier-inhibit means, whereby input to said inhibit means cuts-off said currentamplifier means, whereby thyristor turns on; and, inhibit interface means, whereby desired control-signal input transferred to inhibit means, which said interface means may be chosen from the group comprising: mechanical relay; transformer; and, opto-electric coupler, and share said input signal isolated galvanically from circuit.

85. A logic-compatible relay according to Claim 84 additionally comprising: means limiting current; means limiting voltage; means discerning logic level.

86. A solid-state overvoltage relay according to Claim 20 where said voltage-sensor voltagedropping resistor is the gate-tap means, whereby line-voltage rise above predetermined level will cut off thyristor.

87. A solid-state missing-phase/undervoltage relay according to Claim 20 wherein said thyristor load is a normally-closed mechanical relay supplying a main load with proper number of phases, wherein application of gate shunting will cut off thyristor, not allowing said relay to pull in, whereby power is supplied to said main load, where said line voltage is divided by voltagedivider means and said divided voltage rectified, smoothed, and delayed by capacitor means for each phase independently, said smoothed voltages added to total peak voltage fraction, said total voltage applied to voltage-sensor means which said means isolated galvanically from thyristor, whereby undervoltage caused by missing phase or other cause will halt thyristor shunting, whereby said mechanical relay will energize and puli in, whereby power to main load will be completely cut off.

88. A solid-state missing-phase/undervoltage relay according to Claim 20 wherein said voltage level discerning means comprise the series combination of a suitable zener diode; variable resistance means trip-point control; and, i.e. input to opto-electric coupler, and where latching means may be added.

89. A manual reset latching solid-state electronic fuse according to Claim 20 wherein load current sensed by current-sensor, and resultant voltage-drop rectified, smoothed, and delayed, if necessary, where said diode-junction means may be interfaced with said current-sensor by operational amplifier, where system protected by fast-blow mains fuse, where trip point and time delays may be adjusted, where thyristor turns off immediately in both modes for triac, and where reset is essentially immediate.

90. In a circuit according to Claim 1, the improvement which comprises addition of suitable varistor means across thyristor to protect against line transients.

91. A switching or control circuit according to Claim 1 wherein said apparatus load is a heating; cooling; lighting; inductive; or, mechanical load, or, any combination of same.

92. A d.c. regenerative switch interval/delay relay comprising: source d.c. power; mechanical d.c. relay with N.O. and N.C. contacts; regenerative switch means; rc timing means; reset means; load-connect means; and, interval/ delay selector-switch means, where said regenerative switch means connected in series with said relay and said d.c. power source, where timing capacitor connected across emitter-base of input transistor, where source resistor feeds said capacitor, where relay non-energized until end of timing period, whereby N.C. contacts can carry load current until end said timing period, where at end said timing period said N.O.

contacts can carry load current, where said regenerative switch turns on when voltage across input emitter-base reaches proper value, whereby switch latches on, where reset means turns said regenerative switch off, where total capacitor leakage and said emitter-base current never exceed current supplied by said current source resistor until switching occurs, and where low emitter-base voltage at said switching point ensures minimal source current loss to capacitor.

93. A d.c. solid-state power delay control comprising: source d.c. power; regenerative switch means comprising one power transistor in combination with low-power transistor of opposite polarity; rc timing means; and, reset means, where said regenerative switch connected in series with said power source and load, where timing capacitor connected across emitter-base input transistor, where source resistor feeds said capacitor, where said regenerative switch turns on when voltage across emitter-base reaches proper value, whereby switch latches on, where reset means turn said regenerative switch off, where collector said power transistor connected to power source via proper small-value resistor, where emitter said lower-power transistor member connected via high value resistor to said source, whereby power transistor carries essentially all load current, where total capacitor leakage and said emitter-base current never exceed current supplied by said current source resistor until switching occurs, and where low emitter-base voltage at said switching point ensure minimal source current loss to capacitor.

94. The employment of the special abrupt decrease in emitter-collector resistance of certain thin-base high-frequency Ge transistors such as the OC171 with increased temperature in heating control circuits wherein control current to a control means is shunted by said transistor in lceo move, whereby device controlled by said control means is turned off.

95. A manual reset heating control circuit according to Claim 94 where a triac in a latching circuit, fired in any mode, is cut off when said triac gate shunted by said transistor at transition temperature, said triac remaining off until reset means used.

96. The direct firing of a thyristor gate by the variable-duty-cycle output of an integrated-circuit timer such as of the 555 family, to give a control circuit, whereby load power is varied accordingly.